Deemphasis and subsequent reemphasis of high-energy reversed-spectrum components of a folded video signal

ABSTRACT

A system for reproducing a luminance signal from a medium containing a previously recorded luminance signal with a high-frequency portion thereof compressed in dynamic range includes a circuit for recovering that luminance signal from the medium. Filtering is done to separate the low-frequency and compressed-in-dynamic-range high-frequency portions of the recovered luminance signal from each other. A corer responds to the separated compressed-in-dynamic-range high-frequency portion of the recovered luminance signal to provide a cored high-frequency portion with expanded dynamic range and reduced noise. The level of the cored high-frequency portion with expanded dynamic range and reduced noise is then boosted by a predetermined amount to compensate for energy losses during the compression of the dynamic range and added back to the low-frequency portion of the recovered luminance signal to reproduce a full-band luminance signal with at least partially restored dynamic range for high-frequencies and without readily visible high-frequency noise.

This is a continuation-in-part of U.S. patent application Ser. No.07/604,493 filed Oct. 26, 1990, now abandoned, is a continuation-in-partof U.S. patent application Ser. No. 08/059,765 filed May 11, 1993, nowU.S. Pat. No. 5,500,739, is a continuation-in-part of U.S. patentapplication Ser. No. 07/787,690 filed Nov. 4, 1991, now U.S. Pat. No.6,134,373 and is a continuation-in-part of U.S. patent application Ser.No. 07/635,197 filed Jan. 2, 1991 now abandoned, as acontinuation-in-part of U.S. patent application Ser. No. 07/569,029filed Aug. 17, 1990 and issued May 12, 1992 as U.S. Pat. No. 5,113,262.

The inventions described herein relate to circuitry for deemphasizinghigh frequencies in a luminance signal and for reemphasizing highfrequencies in a reproduced luminance signal, useful in a system fortransmitting a wide bandwidth luminance signal through a narrowbandwidth channel in a backward compatible manner.

RELATED PATENTS & PATENT APPLICATIONS

The subject matter of the present application is related to the subjectmatter disclosed and claimed in the following prior co-pending U.S.patent applications, the claimed inventions of which were commonlyassigned to or under an obligation of assignment to the assignee of thepresent application at the time the respective inventions were made, andin which prior applications at least one named inventor is in commonwith the present application: Ser. No. 531,070 filed May 31, 1990; Ser.No. 545,486 filed Jun. 29, 1990; Ser. No. 599,566 filed Oct. 18, 1990;Ser. No. 604,493 filed Oct. 26, 1990; Ser. No. 607,709 filed Nov. 1,1990; Ser. No. 635,197 filed Jan. 2, 1991; Ser. No. 711,980 filed Jun.7, 1991; Ser. No. 819,890 filed Jan. 13, 1992; Ser. No. 839,542 filedFeb. 24, 1992; Ser. No. 881,131 filed May 11, 1992; Ser. No. 910,491filed Jul. 8, 1992; Ser. No. 996,525 filed Dec. 23, 1992; and Ser. No.059,765 filed May 11, 1993, the disclosures of which prior applicationsare incorporated hereinto by reference thereto.

The specification and drawing of U.S. patent application Ser. No.008,813 filed Jan. 25, 1993 entitled ADAPTIVE DEEMPHASIS AND REEMPHASISOF HIGH FREQUENCIES IN VIDEO TAPE RECORDING, UTILIZING A RECORDEDCONTROL SIGNAL, and assigned to Samsung Electronics Co., Ltd., areappended hereto for purposes of incorporation into this specification.The specification and drawing of U.S. patent application Ser. No.819,890 filed Jan. 13, 1992 entitled DIGITAL MODULATORS FOR USE WITHSUB-NYQUIST SAMPLING OF RASTER-SCANNED SAMPLES OF IMAGE INTENSITY, andassigned to Samsung Electronics Co., Ltd., are appended hereto forpurposes of incorporation into this specification.

The specification and drawings of U.S. patent application Ser. No.08/059,765 filed May 11, 1993 entitled FREQUENCY-MULTIPLEXING FM LUMASIGNAL WITH COLOR AND 2ND UNDER SIGNALS HAVING OVERLAPPING FREQUENCYSPECTRA, and assigned to Samsung Electronics Co., Ltd., are appendedhereto for purposes of incorporation into this specification. Thespecification and drawings of U.S. patent application Ser. No.07/787,690 filed Nov. 4, 1991 entitled SYSTEM FOR RECORDING ANDREPRODUCING A WIDE BANDWIDTH VIDEO SIGNAL VIA A NARROW BANDWIDTH MEDIUM,and assigned to Samsung Electronics Co., Ltd., are appended hereto forpurposes of incorporation into this specification. The specification anddrawings of U.S. patent application Ser. No. 07/604,493 filed Oct. 26,1990 entitled ADAPTIVE DEEMPHASIS AND REEMPHASIS OF HIGH FREQUENCIES INA VIDEO SIGNAL, and assigned to Samsung Electronics Co., Ltd., areappended hereto for purposes of incorporation into this specification.

The specification and drawing of U.S. Pat. No. 5,113,262 issued May 12,1992 to C. H. Strolle et alii and entitled VIDEO SIGNAL RECORDING SYSTEMENABLING LIMITED BANDWIDTH RECORDING AND PLAYBACK are incorporatedherein by reference. The specification and drawing of U.S. Pat. No.5,218,449 issued Jun. 8, 1993 to J. W. Ko et alii and entitled NERVOUSCLOCK SIGNAL GENERATOR FOR VIDEO RECORDER are incorporated herein byreference. The inventions described and claimed in U.S. Pat. Nos.5,113,262 and 5,218,449 were commonly assigned to or under an obligationof assignment to the assignee of the current application at the timesthe respective inventions were made by the current applicants and theirco-inventors.

BACKGROUND OF THE INVENTION

The invention relates generally to a video signal processing system forprocessing a wide bandwidth video signal into a reduced bandwidth signalsuitable for transmission and/or recording via a narrow bandwidth signalmedium, whereby the information content of the wide bandwidth videosignal is retained in the reduced bandwidth signal and the reducedbandwidth signal is compatible with conventional narrow bandwidthreception apparatus, and for receiving and/or reproducing and processingthe transmitted reduced bandwidth signal for recovering therefrom theinformation content of the original wide bandwidth signal. The inventionin certain of its aspects relates more particularly to signal processinguseful in a narrow bandwidth format video cassette recorder (VCR), forconverting a wide bandwidth input video signal to a reduced bandwidthvideo signal containing the information content of the input widebandwidth video signal within the reduced bandwidth, whereby the reducedbandwidth video signal may be recorded and reproduced conventionally bysuch narrow bandwidth format VCR. The invention in other of its aspectsrelates more particularly to processing the reproduced narrow bandwidthvideo signal to recover the information content of the wide bandwidthvideo signal therefrom, whereby a wide bandwidth video signal may bereconstructed for yielding improved video bandwidth of the reproducedsignal comparable to the full bandwidth of the input video signal, whilemaintaining backward compatibility of the recorded reduced bandwidthvideo signal for playing back video cassettes recorded by this improvedvideo signal processing system on available conventional narrowbandwidth format VCRs or video cassette players (VCPs).

Conventional consumer type VCRs record video information onto video tapecassettes in one of several formats. The well-known VHS format systemuses a relatively narrow bandwidth format and produces degraded picturequality in comparison to standard broadcast video chiefly because therecorded VHS format video signal has insufficient horizontal resolution.An enhanced VHS format type recording system, popularly called Super VHSor S-VHS, produces enhanced picture quality by recording a widerbandwidth video signal on the video tape cassette using a higher FMcarrier frequency for the luminance information, thus yielding improvedpicture resolution. Such a format requires a higher FM carrierfrequency, higher quality tape in the cassette and higher qualityrecording and playback mechanisms, heads and circuitry. However, theS-VHS format is not backward compatible with standard VHS format VCRs.I. e., although an S-VHS format VCR can reproduce (playback) cassettesrecorded on either S-VHS format or standard VHS format VCRS, a standardVHS format VCR or VCP cannot play back cassettes recorded on S-VHSformat VCRS.

It is desirable that an improved video recording system be able torecord wider bandwidth video signals on a standard quality cassette thanthose recordable by conventional narrow bandwidth VCRs, while stillmaintaining backward compatibility with conventional narrow bandwidthVCRs, and not require especially high quality magnetic tape or recordand playback mechanisms. That is, it is desirable that normal-quality,narrow-bandwidth recording tape medium video cassettes may be recordablewith wider-bandwidth, higher-frequency video information using theimproved system and be able to be compatibly played back by conventionalnarrow bandwidth VCRs without producing noticeable visual artifacts inthe reproduced image, even if the conventional VCR may not be able toreproduce the full-bandwidth signal recorded on such a cassette.

It has long been a goal of video engineers to increase the amount ofinformation transmitted through a given narrowband channel, such as anNTSC signal channel, which is limited to a nominal 4.2 MHz of usefulbandwidth. Because the frame and line rates (temporal and verticalresolution) usually are fixed, restricting the bandwidth translates intorestricting the horizontal resolution. In some cases, the nominalbandwidth of the channel is limited to 3 MHz or even 2.5 MHz, resultingin an image with insufficient horizontal resolution.

It has long been recognized that in scanned television systems, thesignal energy is concentrated spectrally in the spatio-temporal domainat periodic intervals according to the scanning frequencies, and thevideo spectrum has so-called “holes” or gaps between these discretesignal areas. In these gaps, which gaps also occur at regular intervals,the signal energy is generally comparatively small. The NTSC composite(i. e., “colorplexed”) color video system represents a system which usesone of these “holes” to carry the color information. In the NTSC system,the chrominance or “chroma” signal containing the color information istransmitted combined with the baseband video as a pair ofcolor-difference or mixture signals encoded in quadrature amplitudemodulation of a suppressed nominally 3.58 MHz color subcarrier. I. e.,the color-difference or mixture signals are encoded in respectiveamplitude-modulation sidebands of a pair of in-phase and quadraturecolor subcarriers, both of which subcarriers are suppressed. Thefrequency of the color subcarrier (3.579545 MHz, which is 227.5 timesthe horizontal scanning frequency of 15.734 kHz) was very carefullyselected so that a minimum disturbance occurs when a color video signalis displayed on a black-and-white receiver. Specifically, the NTSC colorsubcarrier frequency is interleaved horizontally, vertically, andtemporally with the luminance signal spectrum to minimize crosstalk andintermodulation between the luminance and chrominance components of thecomposite video signal.

It was recognized at around the time of the adoption of the NTSCcolorplexed system that such frequency spectrum “holes” could also beused to transmit additional horizontal information to increase thehorizontal resolution of the reproduced image. In such systems, thehigher frequency horizontal information is interleaved spectrally withthe lower frequency horizontal information in a similar manner as thechrominance information is in the NTSC color system. An article entitled“REDUCTION OF TELEVISION BANDWIDTH BY FREQUENCY INTERLACE” by E. A.Howson and D. A. Bell in Journal of the British I. R. E., February,1960, pp. 127-136 contains a description of such a system which utilizesanalog signal processing techniques. The artifacts resulting from thefrequency interleaving manifest themselves as annoying dot crawlpatterns in such a system, so it does not accurately reproduce thefull-bandwidth image in its original form.

German Patent Publication No. 82100286.2 entitled “Verfahren zumUbertagen von Fernsehsignalen uber einen genormten bandbreitebegrenztenUbertragunskanal und Anordnung zum Durchfuhren des Verfahrens,” byProfessor Wendland et alii describes principles of offset subsamplingand bandwidth compression as applied to advanced television systems.This Jan. 1, 1982, patent publication also describes techniques forimplementing television systems in accordance with the principlesdescribed therein. Sub-Nyquist sampling is done by replacing every oddsample in a first video line with a zero-valued sample, and then on thenext line, replacing every even sample with a zero-valued sample. Onalternate frames, the patterns are reversed. A television system usingsub-Nyquist sampling at the transmitter can use comb filters in thereceivers to suppress dot crawl patterns arising from spectruminterleaving.

Theoretically, the Howson and Bell frequency folding technique and thesub-Nyquist sampling technique are equivalent, when a folding carrierfrequency 2f_(F) used during spectrum reversal is one-half the samplingfrequency f_(S). The sampled-data digital systems using the sub-Nyquistsampling technique provided improved reconstruction of the receivedimage because of the existence of line and frame combing techniques,which had not been developed at the time of the Howson and Bell system.The sub-Nyquist sampling techniques, however, were developed for totallysampled data digital systems as data reduction (i. e., compression)techniques in digital systems, and the signals generated by thesesampling techniques were riot generally intended to be passed through anarrowband analog channel.

An article, “DEVELOPMENT OF HDTV RECEIVING EQUIPMENT BASED ON BANDCOMPRESSION TECHNIQUE (MUSE)”, by Kojima et alii in IEEE Transactions onConsumer Electronics, Vol. CE-32, No. 4, November 1986, pp.759-768,describes another data compression scheme in which compresses bandwidthby sampling every other pixel each frame, but sampling each pixel onceevery other frame.

This scheme works well only for non-moving images. For moving images, amotion vector is developed, and the actual rate of sampling of eachpixel is adaptively varied in response to the motion vector so that asample of the pixel is transmitted every other frame on the average, butmore often when that pixel is representing a moving image.

U.S. Pat. No. 4,831,463 issued May 16, 1989 to Y. C. Faroudja andentitled VIDEO PROCESSING IN WHICH HIGH FREQUENCY LUMINANCE COMPONENTSARE FOLDED INTO A MIDBAND SPECTRUM describes apparatus for processing avideo signal having a predetermined bandwidth in order to pass the videoinformation contained therewithin through a limited-bandwidth channel,such as magnetic tape. The Faroudja apparatus takes advantage of thefact that the luminance and chrominance components of a composite videosignal are separately recorded on electromagnetic tape in a recordingsystem of the color-under type such as the VHS recording system. In atape recording system of the color-under type the chrominance componentsseparated from the composite video signal are down-converted infrequency to form a color-under signal recorded directly on the tape;and the luminance components separated from the composite video signalare used to modulate the frequency of a luma carrier recorded on thetape as a bias frequency. In the apparatus described in this patent, avideo signal preprocessor includes a comb filter to remove remnantenergy in the spectral holes between spectrally active areas in theluminance signal spectrum that would be occupied by chrominance in theNTSC composite video signal. A folding circuit then folds the basebandvideo luminance signal about a predetermined folding frequency f_(F)selected so that aliases of the baseband luminance signal, generated byheterodyning it with a 2f_(F) folding carrier at twice the frequency ofthe f_(F) folding frequency, are placed into the spectral holes leftafter the chrominance separation and comb filtering. A lowpass filterthen filters the resulting folded video signal so that its bandwidth isabout one-half the bandwidth of the original video signal. The resultingsignal may then be transmitted through the limited-bandwidth channel.

The Faroudja '463 patent further describes a post-processor whichreceives the folded signal from the limited-bandwidth channel. Thepost-processor includes an unfolding circuit which unfolds the receivedsignal about a predetermined unfolding frequency f_(U), which is thesame as the folding frequency f_(F). A comb filter then processes theunfolded signal to remove the alias components resulting from theunfolding process. The signal produced by this comb filter closelyapproximates the original video signal in terms of the bandwidth andinformation content.

It is interesting to note that the Howson and Bell article describes twobandwidth reduction techniques for video luminance signals by frequencyinterlacing or interleaving. In a first technique described, the videoluminance signal spectrum is divided into two equal half-bands (i.e.,band-split at frequency f_(F)), and the upper half-band (i.e., thehighband luminance from frequency f_(F) to frequency 2f_(F)) is used tomodulate a sub-carrier which has its frequency set to be near the upperfrequency limit of the normal video band (i. e., near 2f_(F)). The lowersideband of the modulator output is selected and combined with theoriginal lower half-band. The frequency-interlaced signal resulting fromsuch combining contains all of the original luminance signalinformation, but in one-half the bandwidth of the original signal, andis therefore suitable for transmission over a reduced bandwidth channel.

In a second technique described by Howson and Bell, instead of dividingthe main video luminance signal into two half-bands and modulating the2f_(F) sub-carrier with the high-frequency half-band only, the entiremain video (i. e., baseband) luminance signal is used to modulate the2f_(F) sub-carrier. The lower sideband of the modulator output signalcontains the required interleaved signal in correct frequencyrelationship with the main baseband video signal. If the modulatoroutput is added to the main signal and the resultant added signal ispassed through a lowpass filter having its break frequency atapproximately one-half the sub-carrier frequency, the lowpass filteredoutput signal consists of the correct composite reduced bandwidth signal(with the sub-carrier suppressed). Howson and Bell teach that thissecond technique avoids the need for using complementary lowpass andbandpass filters as required by the first technique employingband-splitting. Howson and Bell adopted this second technique in adescribed experimental apparatus, though the summary abstract appearingin the Howson and Bell article implies that the first technique usingband-splitting was employed.

The folding/unfolding system described in the Faroudja '463 patent issimilar in principle to the second technique described and adopted byHowson and Bell, in the following regards. Howson and Bell selected thefolding modulation sub-carrier frequency to be an odd multiple ofone-half the line scan frequency. In Faroudja '463 the frequency of thefolding heterodyne oscillator/mixer, or of the sub-Nyquist samplingclock applied to the multiplier used as the folding modulator, isselected to be a harmonic of an odd multiple of the line and frame scanrate. Any harmonic of an odd multiple of the line and frame scan rate isan odd multiple of one-half the line scan frequency, supposing there tobe an odd number of scan lines in each frame. In both systems thefolding modulation is performed on the entire baseband luminance signaland lowpass filtering is employed after folding to remove frequenciesgreater than one-half the folding frequency from the folded signal.

In Howson and Bell the high-band luminance is folded into spectral“holes” available because the television signals are luminance-onlysignals for black-and-white television. In Faroudja '463 the high-bandluminance is folded into spectral “holes” from which the NTSCchrominance sub-carrier has been removed. However, because there maystill be residual chroma sidebands present in those areas which mightinterfere with the folding and unfolding processes, the inventors havefound it to be preferable instead to fold the high-band luma into thespectral “holes” described by Fukinuki. Then, any residual chromacomponents when unfolded will be in complementary phase on successivefields and will be optically canceled in the display monitor. In orderto fold the high-band video signal into the same band as the low-bandvideo signal, so as to occupy the Fukinuki portions of the band, thefolding carrier is chosen to be a harmonic of an even multiple of boththe line and the frame scan rates, which harmonic reverses phase fromscan line to scan line and from frame to frame. That is, the phase ofthe folding carrier is reversed at the scan line rate, each reversalbeing at a respective instant between scan lines.

Both the Howson and Bell article and the Faroudja '463 patent describefolding systems which, if incorporated into an improved VCR, producecassettes which cannot be played back on conventional VCRs withoutintroducing unacceptable artifacts into the displayed image. This isprimarily due to the amplitude of the folded high-frequency componentspresent within the spectrum of the low-frequency components on thepreviously recorded cassette. The magnitude of the folded high-frequencycomponents is sufficiently high as to introduce intolerable artifactsand degradation (dot crawl, twinkling, line flicker, etc.) into an imagedisplay produced from a video signal from which the foldedhigh-frequency components are not properly removed.

Howson and Bell were not particularly concerned with backwardcompatibility of the interleaved signal. In fact, they suggestedincluding a pre-emphasis filter for boosting the interleavedhigh-frequency components of the folded luminance signal in order tominimize the effects of crosstalk from the low-frequency luminancecomponents during the transmission of the folded signal through thechannel and to minimize sub-carrier interference at the receiver. If avideo cassette recorded by a VHS format VCR modified to include thesystem taught by Howson and Bell were played back on a standard VHSformat VCR, the pre-emphasized high-frequency components would producean even more objectionable image on the television screen than thatproduced by the Faroudja system.

Thus, the need has remained for improving the video resolution over thatavailable with the currently used, limited-bandwidth video recording andplayback techniques, media and mechanisms in a manner which retainsbackward playback compatibility with existing VCRs and VCPs. Backwardplayback compatibility is greatly improved by lowering the amplitude ofthe high-frequency portion of the luminance signal, before or during itsbeing folded into the low-frequency component, a technique called“deemphasis”. The deemphasized folded luminance signal is then combinedwith the chrominance signal and the combined signal recorded on thevideo tape. Upon playback, the folded deemphasized luminance signal isseparated from the chrominance signal. The separated luminance signal isthen unfolded, and the amplitude of the high-frequency portion isincreased to restore it to substantially its original level, a techniquecalled “reemphasis”. If such a video tape is played back on a VCR whichdoes not have the unfolding and reemphasis circuitry, the artifactscaused by the presence of the high-frequency portion of the luminancesignal within the low-frequency portion can be reduced to levels thatare not objectionable. This is provided for by reducing the amplitude ofthe high-frequency portion of the luminance signal, which causes theartifacts, to a reduced level either prior to the folding procedure orduring the folding procedure.

U.S. Pat. No. 5,113,262 describes a video signal recording system whichincludes an adaptive deemphasis circuit in the luminance signal recordpath and an adaptive reemphasis circuit in the playback path. Theadaptive deemphasis circuit in U.S. Pat. No. 5,113,262 includescircuitry for detecting the level of the high-frequency portion of theluminance signal, and circuitry for variably reducing the level of thehigh-frequency portion in response to the detected signal level. If thelevel of the high-frequency portion of the luminance signal is high,then the level of the high-frequency portion is reduced by a maximumamount; if the level is low, then the level is reduced by a minimumamount. This operation can be referred to as “compression of the dynamicrange of luminance signal high frequencies”.

A problem that has subsequently been noted by the inventors with regardto the adaptive deemphasis circuit described in U.S. Pat. No. 5,113,262arises when recording composite video signals that have accompanyingnoise in the upper luminance frequencies. The circuitry for detectingthe level of the high-frequency portion of the luminance signal detectsthe noise is detected, and the level of the high-frequency portion isreduced in response to the detected noise. This undesirably wipes outlow-level high-frequency luminance detail, causing textured surfaces toappear smooth. This problem is solved in an improved adaptive deemphasiscircuit that embodies an aspect of the invention. In this improvement athreshold circuit is introduced after the detector that detects theamplitude of the high-frequency portion of the luminance signalcomponent. This threshold circuit operates as a corer, or base-lineclipper. Accordingly, the high-frequency portion of the luminance signalis not deemphasized until its amplitude exceeds a threshold level. Thisthreshold level is set to be above the level of noise in thehigh-frequency portion of the luminance signal component of thecomposite video signal received for recording.

An adaptive deemphasis circuit and control signal generator constructedin accordance with this aspect of the invention can also be useful inimplementing the video tape recording and playback systems described inU.S. patent application Ser. No. 008,813 filed Jan. 25, 1993 byChristopher H. Strolle and Raymond A. Schnitzler, and entitled ADAPTIVEDEEMPHASIS AND REEMPHASIS OF HIGH FREQUENCIES IN VIDEO TAPE RECORDING,UTILIZING A RECORDED CONTROL SIGNAL. U.S. patent application Ser. No.008,813 is a continuation-in-part of U.S. patent application Ser. No.604,494 filed Oct. 26, 1990 by Strolle and Schnitzler, and entitledADAPTIVE DEEMPHASIS AND REEMPHASIS OF HIGH FREQUENCIES IN A VIDEO SIGNALUTILIZING A RECORDED CONTROL SIGNAL. The inventions described andclaimed in both these Strolle and Schnitzler applications were commonlyassigned to or under an obligation of assignment to the assignee of thecurrent application at the times the respective inventions were made bythe current applicants and their co-inventors.

The adaptive reemphasis circuit in the playback path of U.S. Pat. No.5,113,262 performs an operation which can be referred to as “expansionof the dynamic range of luminance signal high frequencies”, which issubstantially the inverse of the “compression of the dynamic range ofluminance signal high frequencies” operation carried out by the adaptivedeemphasis circuit during recording. The adaptive deemphasis andadaptive remphasis procedures described in U.S. Pat. No. 5,113,262 arecarried out in response to the high-frequency content of the luminancesignal in static as well as dynamic portions of the image. The adaptivereemphasis circuit in the playback path includes circuitry for detectingthe level of the high-frequency portion of the unfolded luminancesignal, and circuitry for variably increasing the level of thehigh-frequency portion in response to the detected level. If the levelof the high-frequency portion of the unfolded luminance signal isrelatively high, then the level is boosted by the maximum amount; if thelevel is relatively low, then the level is boosted by the minimumamount.

Because when the level of the high-frequency portion of the luminancesignal is high it is reduced by a maximum amount and when it is low itis reduced by a minimum amount, the level of the high-frequency portionis controlled to always be at about the same level. This deemphasizedluminance signal is then folded, recorded, played back and unfolded.Each of these steps may introduce noise.

It is desirable to produce the highest quality image possible in a VCRwith unfolding circuitry in the playback path, while maintainingbackward compatibility with VCRs without the unfolding circuitry in theplayback path. The compression of the dynamic range of luminance signalhigh frequencies done before recording a video tape should becompensated for by a complementary expansion of the dynamic range ofluminance signal high frequencies reproduced from the video tape. Thisrequires that the amount by which the high-frequency portion of theluminance signal is reduced in the record path substantially correspondsto the amount by which the high-frequency portion of the luminancesignal is increased in the playback path. A solution described in U.S.Pat. No. 5,113,262, includes an adaptive reemphasis circuit in theplayback path. The level of the high-frequency portion of the reproducedluminance after its unfolding is detected and used to control thehigh-frequency gain of the adaptive reemphasis circuit.

This circuit at times responds unfavorably to noise introduced in therecord and playback process. If the introduced noise changes thedetected level of the high-frequency portion, the reemphasis function isno longer substantially the inverse of the deemphasis function over theentire dynamic range of luminance high frequencies. Solutions to thisproblem are provided by reemphasis circuits that are constructed inaccordance with aspects of the present invention.

Another solution to this problem is described in described in U.S.patent application Ser. No. 604,494 filed Jan. 25, 1993. A controlsignal, which is generated for controlling the deemphasis of thehigh-frequency portion of the luminance signal during its processingbefore being recorded, is encoded for recording on the video tape.During playback from the video tape, the control signal is reproducedand used for controlling the reemphasis of the high-frequency portion ofthe luminance signal. This avoids adaptive reemphasis circuitry thatresponds unfavorably to noise introduced in the record and playbackprocess, but complicates the recording and playback proceduresconsiderably.

Recording and playback procedures for transmitting auxiliary signals,such as a control signal for controlling the reemphasis of thehigh-frequency portion of the luminance signal during playback, aredescribed in U.S. patent application Ser. No. 059,765 filed May 11, 1993by Christopher H. Strolle, Werner F. Wedam, Jung-Wan Ko, Chandrakant B.Patel and Allen L. Limberg, entitled FREQUENCY-MULTIPLEXING FM LUMASIGNAL WITH COLOR AND SECOND UNDER SIGNALS HAVING OVERLAPPING FREQUENCYSPECTRA, and assigned to Samsung Electronics Co., Ltd., pursuant toobligations of the inventors to so assign their invention at the time ofits making. U.S. patent application Ser. No. 027,772 is acontinuation-in-part of their U.S. patent application Ser. No. 531,070filed May 31, 1990, entitled CHROMA CHANNEL ENCODED WITH AUXILIARYSIGNALS, and assigned to Samsung Electronics Co., Ltd., pursuant toobligations of the inventors to so assign their invention at the time ofits making.

Accordingly, an improvement of the adaptive reemphasis circuitrydescribed in U.S. Pat. No. 5,113,262 is desirable to have, whichimprovement does not respond unfavorably to noise introduced in therecord and playback process, and which improvement does not require acontrol signal to be encoded for recording on the video tape or to bedecoded when playing back from the video tape.

SUMMARY OF THE INVENTION

A first aspect of the invention concerns an adaptive deemphasis circuitfor reducing the amplitude of the high-frequency portion of a luminancesignal respective to its low-frequency portion, for producing a backwardcompatible video tape recording wherein the high-frequency portion ofthe luminance signal is folded into its low-frequency portion to form afolded-spectrum luminance signal for modulating the frequency of aluminance carrier wave.

In accordance with a second aspect of the invention, an adaptivedeemphasis circuit embodying the first aspect of the invention can beimproved so as to be non-responsive to noise in the high-frequencyportion of the luminance signal component of the composite video signalreceived for recording. In this improvement a threshold circuit isintroduced after the detector that detects the amplitude of thehigh-frequency portion of the luminance signal component. This thresholdcircuit operates as a corer, or base-line clipper. Accordingly, thehigh-frequency portion of the luminance signal is not deemphasized untilits amplitude exceeds a threshold level which is above the level ofnoise in the high-frequency portion of the luminance signal component ofthe composite video signal received for recording.

A third aspect of the invention concerns adaptive reemphasis circuitsfor reproducing a luminance signal from a medium containing a previouslyrecorded luminance signal with its high-frequency portion compressed indynamic range and folded into its low-frequency portion.

A fourth aspect of the invention concerns adaptive reemphasis circuitsembodying the third aspect of the invention and avoiding the emphasizingof noise during reemphasis of the high-frequency portion of theluminance signal compressed in dynamic range during recording. Filteringis done to separate the low-frequency portion of the recovered luminancesignal and its compressed-in-dynamic-range high-frequency portion fromeach other. A corer responds to the separatedcompressed-in-dynamic-range high-frequency portion of the recoveredluminance signal to provide a cored high-frequency portion with expandeddynamic range and reduced noise. The level of the cored high-frequencyportion with expanded dynamic range and reduced noise is then boosted bya predetermined amount to compensate for energy losses in the coringprocedure and added back to the low-frequency portion of the recoveredluminance signal to reproduce a full-band luminance signal with at leastpartially restored dynamic range for high-frequencies and withoutreadily visible high-frequency noise.

Further aspects of the invention concern the preferred choice of thefolding carrier frequency to be a harmonic of an even multiple of boththe line and the frame scan rates, which harmonic is then reversed inphase at the scan line rate, each reversal being at a respective instantbetween scan lines, to form the folding carrier. Such folding carrierconditions the folding circuitry for folding a high-band luminancesignal into the Fukinuki holes of the low-band luminance signal duringrecording. A corresponding unfolding carrier is used during playback forunfolding the high-band luminance signal from the Fukinuki holes of thelow-band luminance signal, after which the low-band and high-bandluminance signals are combined to recover a full-band luminance signal.

BRIEF DESCRIPTION OF THE DRAWING

FIG. 1 is a block diagram of a portion of a video signalrecording/playback system constructed in accordance with principles ofthe present invention.

FIG. 2 is a block diagram of an encoder used in the recording portion ofthe FIG. 1 system.

FIG. 3 is a plot of the vertical-spatial-frequency versushorizontal-spatial-frequency characteristics of a folding modulationperformed in the folding circuit.

FIG. 4 is a plot of the vertical-spatial-frequency versustemporal-frequency characteristics of the same folding modulation.

FIG. 5 is a block diagram showing in more detail an adaptive luminanceseparation portion, a motion signal generating portion and a chrominancesignal separating portion of the FIG. 2 encoder.

FIG. 6 is a block diagram showing an implementation of the adaptiveluminance separation portion and motion signal generating portion of theFIG. 5 encoder.

FIG. 7 is a block diagram showing in more detail a soft switch employedin an adaptive luminance filtering section of the FIG. 2 encoder.

FIG. 8 is a detailed block diagram of a chrominance/motion combiningcircuit in the FIG. 2 encoder.

FIG. 9 is a block diagram of a decoder used in the playback portion ofthe FIG. 1 system.

FIG. 10 is a block diagram of unfolding circuitry used in the FIG. 9decoder.

FIG. 11 is a block diagram showing in more detail a soft switch employedin a spatio-temporal post filter circuit of the FIG. 10 unfoldingcircuitry.

FIG. 12 is a detailed block diagram of chrominance/motion signalseparating circuitry in the FIG. 9 decoder.

FIG. 13 shows the coefficients of a digitally implemented quadrantselective filter in the FIG. 12 chrominance/motion signal separatingcircuitry.

FIG. 14 is a block diagram of circuitry for generating a folding carrierof a frequency that is a multiple of scan line frequency, the phase ofwhich folding carrier is reversed between each pair of successive scanlines.

FIG. 15 is a block diagram of folding circuitry that can be used afteradaptive deemphasis circuitry which in accordance with the inventioncompresses the high-frequency luminance signal.

FIG. 16 is a block diagram of one type of adaptive deemphasis circuitwith control signal generator, which adaptive deemphasis circuitembodies an aspect of the invention.

FIG. 17 is a plot of amplitude versus normalized frequency for afull-bandwidth input luminance signal supplied for folding.

FIG. 18 is a plot of amplitude versus normalized frequency of theluminance signal as folded to half bandwidth by sub-Nyquist samplingprocedure.

FIG. 19 is a plot of amplitude versus normalized frequency of theluminance signal as folded to half bandwidth by sub-Nyquist samplingprocedure with fixed deemphasis of the high-frequency component.

FIG. 20 is a plot of amplitude versus normalized frequency of theluminance signal as folded to half bandwidth by sub-Nyquist samplingprocedure with adaptive deemphasis of the high-frequency component inaccordance with an aspect of the invention.

FIG. 21 is a plot of amplitude versus normalized frequency of theluminance signal as folded to half bandwidth by sub-Nyquist samplingprocedure with adaptive deemphasis of the high-frequency component andwith noise coring.

FIGS. 22, 23 and 24 are block diagrams of alternative ways ofimplementing the combination of lowpass filter and highpass filter usedin the adaptive deemphasis and reemphasis circuits embodying aspects ofthe invention.

FIG. 25 is a block diagram of a particular adaptive deemphasis circuitof the type shown in FIG. 16, which FIG. 25 circuit embodies an aspectof the invention and can be used in the video signal recording/playbacksystem illustrated in FIG. 1.

FIG. 26 is a set of related waveform diagrams which are useful inunderstanding the operation of the FIG. 25 adaptive deemphasis circuit.

Each of FIGS. 27, 28 and 29 is a block diagram of an adaptive reemphasiscircuit which embodies the invention in one of its aspects and can beused in conjunction with the FIG. 25 deemphasis circuit.

FIG. 30 is a block diagram of a control signal generator including aread-only memory (ROM) for storing a look-up table of control signalvalues, which control signal generator can be used in the FIG. 16 typeof adaptive deemphasis circuit in accordance with an aspect of theinvention.

FIG. 31 is a plot of the control signal function provided by the look-uptable of control signal values of the FIG. 30 control signal generatorROM, against the level of the energy of those luminance highfrequencies; and is further an additional plot of the resultantdeemphasis of the luminance high frequencies with noise coring, plottedagainst the same abscissa.

FIG. 32 is a block diagram of an adaptive reemphasis circuit constructedin accordance with an aspect of the invention, which adaptive reemphasiscircuit includes a control signal generator that has therewithin aread-only memory (ROM) for storing a look-up table of control signalvalues.

FIG. 33 is a plot of a control signal function as may be provided by thelook-up table of control signal values of the FIG. 32 control signalgenerator ROM, plotted against the level of the energy of thoseluminance high frequencies; and is further an additional plot of theresultant deemphasis of the luminance high frequencies with noisecoring, plotted against the same abscissa.

FIG. 34 is a block diagram of combined adaptive deemphasis and foldingcircuitry, as embodies another aspect of the invention.

FIG. 35 is a detailed block diagram of a folding modulator in the FIG.34 combined adaptive deemphasis and folding circuitry.

FIG. 36 is a block diagram of a modification of the FIG. 34 combinedadaptive deemphasis and folding circuitry, which modified combinedadaptive deemphasis and folding circuitry embodies an aspect of theinvention.

FIGS. 37 and 38 are block diagrams of still further alternativeimplementations of adaptive deemphasis and folding circuits inaccordance with aspects of the invention.

DETAILED DESCRIPTION OF THE INVENTION

A system embodying the present invention in its various aspects may beimplemented using analog and/or digital signal processing techniques.For sake of example, an implementation of the system embodying thepresent invention in its various aspects will be described below usingdigital signal processing. However, given the description herein, one ofordinary skill in the art will understand that the system embodying thepresent invention in various of its aspects may be constructed usinganalog circuitry and analog signal techniques and how such may beimplemented.

Equalizing delays have been omitted from the drawing figures to simplifythem and to make them easier to understand. One skilled in the art ofvideo signal processor design will appreciate the need for such delaysto properly time-align pixels subject to different delays on differentprocessing paths due to the differing processing performed in thosepaths. One skilled in the art would understand where such delays wouldbe necessary and how long each of the delays would have to be, and suchdelays will not be described or discussed below.

Additionally, various filters are shown in the drawing FIGS. forfiltering in the horizontal, vertical, and temporal directions, havingboth highpass and lowpass response characteristics. One skilled in theart of video signal processor disign will appreciate that some of suchfilters may be constructed as known tapped-delay-line filter or combfilter designs, and would understand how to properly select the delayperiods of the respective delay lines, the number of taps and theweighting of the taps. Consequently, the detailed design of suchtapped-delay-line filters and comb filters will not be discussed below,unless such a design detail is important for other reasons.

In logic circuitry, such as that used in implementing the generation offolding carrier, or such as that used in implementing the generation ofunfolding carrier, one skilled in the art would understand how toprovide the timing delays required to overcome undesired “logic race”conditions; and the detailed design of such logic circuitry to forestallsuch undesired “logic race” conditions will not be discussed below.

Further, where an analog-to-digital converter (ADC) is shown ordescribed in the present disclosure, one skilled in the art wouldunderstand the desirability of preceding such converter with ananti-aliasing lowpass filter, and how this could be implemented, andsuch will not be further described in detail below. Also, where adigital-to-analog converter (DAC) is shown or described in the presentdisclosure, one skilled in the art would understand the desirability offollowing such converter with a sampling clock rejection lowpass filter,and how this could be implemented, and such will not be furtherdescribed in detail below.

Also, in the drawing and in the following detailed description, variousembodiments constructed in accordance with the present invention itsvarious aspects are directed to an NTSC composite video baseband signal.One skilled in the art would understand how to modify the embodiments inorder to process a PAL video signal, a SECAM video signal or a videosignal according to any other standard in accordance with the teachingof the present inventors set forth herein.

In the FIG. 1 video signal recording/playback system, a luminance signalinput terminal 5 is supplied a full-bandwidth luminance signal, from theluminance output terminal of a luminance-chrominance separator in thevideo signal recording/playback system, or from the luminance outputterminal of a video camera, by way of example. The luminance signalinput terminal 5 connects to a signal input terminal of an adaptivedeemphasis circuit 10 and to an input terminal of a control signalgenerator 80. A control signal output terminal of the control signalgenerator 80 connects to a control signal input terminal of the adaptivedeemphasis circuit 10. An output terminal of the adaptive deemphasiscircuit 10 connects to an input terminal of a folding circuit 20. Anoutput terminal of the folding circuit 20 connects to a luminance signalinput terminal of combining circuitry 30. A chrominance signal inputterminal 25 is supplied a chrominance signal, from the chrominanceoutput terminal of the luminance-chrominance separator in the videosignal recording/playback system, or from the chrominance outputterminal of the video camera, with regard to the previously citedexamples. Chrominance signal input terminal 25 connects to a chrominancesignal input terminal of the combining circuitry 30. An output terminalof the combining circuitry 30 supplies a combined signal to a mechanismfor recording the combined signal on a recording medium. The recordingmechanism and the medium are made up of well known elements.

In FIG. 1 a video cassette recorder 40 is a representative recordingmechanism, which uses video tape cassettes as the recording medium. Thecombining circuitry 30 that precedes the VCR 40 will then include anoscillator for generating a luminance carrier wave, circuitry formodulating the frequency of the luminance carrier in accordance with thefolded luminance signal, circuitry for encoding the chrominance signalin an under signal lower in frequency than the frequency-modulatedluminance carrier, and a frequency multiplexer for combining thefrequency-modulated luminance carrier with the under signal.

A video cassette player 45 comprises the known elements making up theplayback mechanism for retrieving the previously recorded signal fromthe recording medium. (The recorder 40 or the player 45 or each of themmay, in fact, be a machine including a video cassette transportmechanism with both recording and playback electronics.) Signal playedback from the video cassette 45 is supplied to an input terminal of asignal separator 50. A luminance signal output terminal of signalseparator 50 connects to an input terminal of an unfolding circuit 60.An output terminal of the unfolding circuit 60 connects to a signalinput terminal of a reemphasis circuit 70. An output terminal of thereemphasis circuit 70 supplies reconstructed full-bandwidth luminancesignal to luminance signal output terminal 15, for subsequentapplication to means (not shown) for utilizing the reconstructedfull-bandwidth luminance signal. That means may be aluminance-chrominance signal combiner for generating a composite videosignal, or that means may be the luminance signal input terminal of ahigh-resolution television monitor, by way of example.

A chrominance signal output terminal of the separating circuit 50connects to a chrominance signal output terminal 35, for subsequentapplication to means (not shown) for utilizing the chrominance signal.By way of example, that means may be the luminance-chrominance signalcombiner for generating a composite video signal or may be thechrominance signal input terminal of the high-resolution televisionmonitor.

One familiar with video signal recording/playback systems willunderstand that elements other than those illustrated in FIG. 1 must bepresent in a recording/playback system. One skilled in the art wouldunderstand where these elements should be placed and how they should beinterconnected. For clarity, these elements have been omitted from FIG.1, and will not be discussed in detail below.

In operation, the control signal generator 80 produces a control signalwhich represents the level of the high-frequency portion of thefull-bandwidth luminance signal. The control signal is applied to thecontrol signal input terminal of the adaptive deemphasis circuit 10. Theadaptive deemphasis circuit 10 operates to variably decrease the levelof the high-frequency portion of the full-bandwidth video signal inresponse to the control signal from the control signal generator 80. Theadaptive deemphasis circuit 10 and control signal generator 80 aredescribed in detail below.

The high-frequency portion of the deemphasized luminance signal from theadaptive deemphasis circuit 10 is then folded into the low-frequencyportion in the folding circuit 20. The folded deemphasized luminancesignal from the folding circuit 20 and the chrominance signal from thechrominance signal input terminal 25 are combined in the combiningcircuitry 30 to form a frequency-division-multiplex signal forrecording. The chrominance signal is down-converted in frequency to forma color-under signal recorded in a spectral region below the spectralregion in which a luma carrier having its frequency modulated inaccordance with the folded deemphasized luminance signal is recorded;and the frequency-division-multiplex signal combining the color-undersignal and the frequency-modulated luma carrier occupies a bandwidthwhich is less than the bandwidth of the magnetic medium.

Upon playback, the reproduced frequency-division-multiplex signal fromvideo cassette player 45 is processed by a separating circuit 50 in aknown manner. In separating circuit 50, the color-under signal and thefrequency-modulated luma carrier are separated by frequency-selectivefiltering, the frequency modulation of the luma carrier is detected torecover the folded deemphasized luminance signal, and the color-undersignal is upconverted to regenerate the chrominance signal. The playedback folded luminance signal is supplied to the unfolding circuit andthe chrominance signal is supplied to the chrominance signal outputterminal 35.

The unfolding circuit 60 unfolds the deemphasized high-frequency portionof the luminance signal from the low-frequency portion, and regeneratesthe deemphasized full-bandwidth luminance signal. This unfoldeddeemphasized full-bandwidth luminance signal is supplied to the signalinput terminal of the reemphasis circuit 70. The reemphasis circuit 70boosts the high-frequency portion of the luminance signal by an amountdependent on its energy. The output of the reemphasis circuit 70 is afull-bandwidth luminance signal in which the high-frequency portion hasbeen restored to substantially the correct level.

Because the high-frequency portion of the luminance signal is attenuatedbefore it is folded into the low-frequency portion in folding circuit20, when the thus recorded video cassette is subsequently played back ona VCR which does not have the unfolding circuit, the artifacts caused bythe presence of the high-frequency portion are not objectionable. Such acassette is backward compatible.

As is described more fully further on in this specification, incombining circuitry 30 of a preferred type for the FIG. 1 video signalrecording/playback system, a motion representative signal M is developedby analysis of the frame-to-frame change in the video luminancecomponents of the television signal supplied for recording, and thismotion representative signal M is utilized as a control signal formotion-adaptive processing of the video luminance components of theinput video signal prior to folding. During reproduction or playback,use of the same motion representative signal M utilized during therecord side luminance processing can significantly facilitate luminanceprocessing during reconstruction, so the motion representative signal Mis additionally processed in combining circuitry 30 to advantageouslycombine it with the video chrominance component signal C to provide acomposite chrominance-plus-motion signal (C+M)_(R) to befrequency-multiplexed with the folded luminance signal Y_(R), thereby togenerate the frequency-multiplexed signal supplied to the VCR 40 forrecording. Details of the encoding and decoding of the motionrepresentative signal M will be more fully described later. Thechrominance-plus-motion signal (C+M)_(R) is recorded as a compositeunder signal instead of the conventional color-under signal recorded ina standard VCR. In a VHS format VCR for example, the 3.58 MHz NTSCchrominance sub-carrier frequency is heterodyned with a 4.21 MHz carrierto down-convert it to about 629 kHz to provide a color-under carrier. Inaccordance with an aspect of the present invention, the compositechrominance-plus- motion signal (C+M)_(R) is modulated on another undercarrier and combined with the luminance carrier wave frequency-modulatedby the luminance signal Y_(R). The VCR 40 of FIG. 1 records theresulting frequency-multiplexed signal onto a video tape cassette inconventional manner, substantially in accordance with the standard VHSVCR format.

In connection with the modified chrominance encoding and decoding, thoseskilled in the art will understand that the NTSC chroma carrier is atwo-line sequence, out of phase by 180° at the same horizontal positionalong adjacent lines, whereas the color-under signal carrier is afour-line sequence advanced or retarded in phase by 90° per line at thesame horizontal position, with the phase advanced by 90° per line orretarded by 90° per line on alternate field tracks. The other undercarrier is a four-line sequence, with the phase retarded by 90° per lineduring the field tracks the color-under signal carrier is advanced by90° per line, and the phase advanced by 90° per line during the fieldtracks the color-under signal carrier is retarded by 90° per line.

FIG. 2 is block diagram showing in more detail the recording circuitrypreceding the VCR 40. For facilitating the signal processing, digitalsignal processing techniques may advantageously be implemented, and soinput terminal 105 is coupled to an input terminal of ananalog-to-digital converter (ADC) 102 which produces a digitized(quantized) composite video output signal V responsive to the analogcomposite video signal received for recording. An output terminal of ADC102 is coupled to respective input terminals of an adaptive luminancesignal separator 104, a motion signal generator 106 and a chrominancesignal separator 114. An output terminal of the adaptive luminancesignal separator 104 is coupled to an input terminal of deemphasis andfolding circuitry 108. As will be more fully described below, thedeemphasis and folding circuitry 108 performs band-splitting of theseparated luminance signal from luminance separator 104 intolow-frequency and high-frequency luminance components, performs anadaptive deemphasis processing of the high-frequency luminancecomponent, and folds the adaptively deemphasized high-frequencyluminance component into the low-frequency luminance component spectrum,thereby to provide a digital bandwidth-limited folded luminance signalY_(F).

An output terminal of the deemphasis and folding circuitry 108 iscoupled to an input terminal of a digital-to-analog converter (DAC) 110by which the digital folded luminance signal Y_(F) is converted to ananalog signal Y_(R). The signal Y_(R) is supplied to avoltage-controlled oscillator (VCO) 120 for modulating the frequency ofa luminance carrier wave. The frequency-modulated luminance carrier waveis supplied from an output terminal of the VCO 120 to the input terminalof a bandpass filter 121, which constrains the bandwidth of thefrequency-modulated oscillations so their frequency spectrum does notoverlap the frequency spectrum of the color-under signal or thefrequency spectrum of a lower-carrier-frequency second under signal. Theband-limited frequency-modulated oscillations are applied as a firstsummand input signal to an analog adder 122, to be combined with undersignal(s) in the adder 122 to generate a sum output signal supplied atthe terminal 124, which is the output terminal of the combiningcircuitry 30.

A separated chrominance signal output terminal of the chrominance signalseparator 114 is coupled to a chrominance signal input terminal of achrominance/motion signal combining circuit 116. A motion representativesignal output terminal of the motion signal generator 106 is coupled toa control input terminal of the adaptive luminance signal separator 104and a motion signal input terminal of the chrominance/motion signalcombining circuit 116. The chrominance/motion signal combining circuitincludes downconverters that generate a composite digitalchrominance-plus-motion signal (C+M) output that is lower in frequencythan the frequency-modulated luminance carrier wave supplied from thecontrolled oscillator 120. The composite digital chrominance-plus-motionsignal (C+M) output by chrominance/motion signal combining circuit 116is coupled to an input terminal of a second digital-to-analog converter(DAC) 118 which supplies an analog chrominance-plus-motion signal(C+M)_(R). An output terminal of DAC 118 supplies a second summand inputsignal to the analog adder 122, to provide an under signal forfrequency-division-multiplexing with the frequency-modulated luminancecarrier wave from the controlled oscillator 120, to generate the sumoutput signal the adder 122 supplies at the terminal 124.

In operation, initially the ADC 102 in the FIG. 2 circuitry converts thecomposite video signal input at input terminal 105 to a sampled datamulti-bit digital composite video signal V. For an NTSC signal having anominal bandwidth extending from DC up to approximately 4.2 MHz, forexample, the sampling frequency may be selected to be about 10 MHz.Digital composite video signal V is supplied to the adaptive luminanceseparator 104, which extracts the luminance component Y therefrom; tothe motion signal generator 106, which derives therefrom a motionrepresentative signal M (hereafter simply referred to as “motion signalM”) for controlling the motion-adaptive filtering on the encoder sideand also on the decoder side; and to the chrominance signal separator114, which extracts the chrominance component C therefrom. Thischrominance component C comprises quadrature-amplitude-modulation (QAM)sidebands of the suppressed color subcarrier, which subcarrier is at3.58 MHz for television signals adhering to the NTSC standard.

As controlled by motion signal M, the adaptive luminance separator 104performs a motion-adaptive spatio-temporal filtering of the digitalcomposite video signal V, to separate the luminance signal Y. It isknown in the video signal processing art that frame-comb lowpassfiltering (temporal lowpass filtering) may be used to extract theluminance component from a composite video signal with no loss ofspatial resolution. However, in the presence of motion in the videoimage, undesirable artifacts are introduced into the luminance signalextracted by frame-comb filtering. Line-comb lowpass filtering (verticalcomb lowpass filtering or spatial lowpass filtering) may also be used toextract the luminance component, even in the presence of motion.However, the luminance component extracted by line-combing has decreasedspatial (diagonal) resolution. It is preferable to extract the luminancesignal using frame-comb filtering in order to preserve spatialresolution, unless there is motion in an area of the image, in whichcase it is preferable to use line-comb filtering in that area.

The extracted luminance signal Y is further processed by the adaptivedeemphasis and folding circuitry 108. This circuit folds the adaptivelydeemphasized high-frequency component of the luminance signal Y into thebandwidth of the lower frequency luminance component so that all theinformation in the full-bandwidth baseband luminance signal Y iscontained in a folded luminance signal Y_(F) having a reducedbandwidth—extending up from DC only to about 2.5 MHz, for example. Theadaptive deemphasis and folding circuitry 108 will be described in moredetail below. The folded luminance signal Y_(F) is converted to ananalog signal Y_(R) in the DAC 110. The signal Y_(R) modulates thefrequency of the controlled oscillator 120.

In the chrominance/motion signal combining circuitry 116, the extractedmotion signal M is used to modulate a carrier. That modulated carrierand the extracted chrominance component signal C are combined into asingle composite chrominance-plus-motion signal (C+M) supplied from thecircuitry 116 to the DAC 118. A chrominance/auxiliary signal combiningcircuit, which may be used as the chrominance/motion signal combiningcircuit 116, is described in more detail in prior copending commonlyassigned U.S. patent application Ser. No. 531,070 filed May 11, 1993 byC. H. Strolle et alii, entitled FREQUENCY MULTIPLEXING FM LUMA SIGNALWITH COLOR AND SECOND UNDER SIGNALS HAVING OVERLAPPING FREQUENCY SPECTRAand assigned to Samsung Electronics Co., Ltd., pursuant to theobligations of the inventors to so assign their inventions at the timethose inventions were made.

The chrominance-plus-motion signal (C+M) is converted into an analogsignal (C+M)R by DAC 118. This (C+M)_(R) signal is in a form which canbe combined in the adder 122 with frequency-modulated luminance carrierwave from the controlled oscillator 120, (C+M)_(R) being an under signalin a frequency-division-multiplex (FDM) signal supplied at the terminal124, which the combining circuitry 30 supplies in FIG. 1 to the VCR 40for recording.

It is pointed out that because the folded highs alternate in phase at 15Hz, it is impractical to detect frame-to-frame motion after folding theluminance signal frequency spectrum. Accordingly, when recording, motionis detected prior to folding. This is done by temporal differencing andspatial lowpass filtering of the separated baseband luminance prior tofolding.

It is helpful for an understanding of certain aspects of the inventionto provide a further explanation regarding the choice of the folding andprefiltering processing employed in the adaptive deemphasis and foldingcircuitry 108, so as to be able to record and reproduce a full-bandwidth(e. g., DC to 5 MHz) luminance signal using a narrow bandwidth videorecording format such as the conventional VHS format. It has previouslybeen proposed to shift a high-frequency luminance signal component byfiltering and sub-Nyquist sampling to insert it within spectral holeswithin the spatio-temporal frequency domain occupied by the NTSCchrominance sub-carrier component, but offset relative thereto. See forexample, T. Fukinuki et alii, “Extended Definition TV Fully Compatiblewith Existing Standards”, IEEE Transactions on Communications, Vol.COM-32, No. 8, August 1984, pages 948-953; and T. Fukinuki et alii,“NTSC FULL COMPATIBLE EXTENDED DEFINITION TV PROTO MODEL AND MOTIONADAPTIVE PROCESSING”, reprinted from IEEE Communications Society “IEEEGlobal Telecommunications Conference”, No. 4.6, Dec. 2-5, 1985, pages113-117; the disclosures of which are incorporated hereinto by referencethereto.

FIGS. 3 and 4 show the frequency characteristics of the folding processemployed in an aspect of the invention, in the vertical-horizontalfrequency spectrum and the vertical frequency-temporal domains,respectively. The folding frequency f_(F) is selected so as to maximizethe distance between the frequency of the folding carrier and thebaseband luminance signal in the temporal, vertical and horizontaldirections. As illustrated in FIGS. 3 and 4, the folding carrier ispreferably placed at one-half the maximum vertical frequency, andone-half the maximum temporal (frame) frequency, to correspond to theso-called Fukinuki holes in the temporal and vertical dimensions, and ata 2f_(F) of about 5 MHz in the horizontal direction. This maximizes thespectral distance between the folding carrier and the vertical andtemporal lower frequency components of the luminance signal. Thehigh-band luminance is folded into the so-called “Fukinuki” areas in theupper left and lower right quadrants of the diamond in FIG. 4. Becauseconventional VCRs employ a component type recording system, it wouldalso have been possible to fold the high-band luminance into thespectral “holes” from which the NTSC chrominance sub-carrier has beenremoved in the upper left and lower right quadrants of the diamond.However, because there may still be residual chroma sidebands present inthose areas which might interfere with the folding and unfoldingprocesses, the inventors have found it preferable to fold the high-bandluma into the Fukinuki areas as shown, with the result that by foldinginto these quadrants, any residual chroma components when unfolded willbe in complementary phase on successive fields and will be opticallycanceled in the display monitor.

Preferably, then, the folding carrier is not a harmonic of an oddmultiple of both the line and the frame scan rates, which harmonic iscontinuous in phase from scan line to scan line and from frame to frame,as described by Howson and Bell and later by Faroudja. Preferably, thefolding carrier is chosen to be a harmonic of an even multiple of boththe line and the frame scan rates, which harmonic reverses phase fromscan line to scan line. i.e., the harmonic reverses phase from scan lineto scan line within each frame and from frame to frame. When thislatter, preferred type of folding carrier is multiplied by a modulatingsignal of lower frequency, a form of amplitude modulation referred to as“4-field offset modulation” takes place. Any harmonic of an evenmultiple of the line and frame scan rate is not an odd multiple ofone-half the scan line frequency; but, rather, is an even multiple ofone-half the scan line frequency—which is to say, a multiple of the scanline frequency.

The generation of a folding carrier that is a harmonic of the scan linefrequency is readily accomplished using a line-locked voltage-controlledoscillator (VCO) that oscillates at an even multiple of the foldingcarrier and frequency-dividing from those oscillations to derive thefolding carrier. Automatic frequency and phase control (AFPC) voltagefor controlling the VCO is developed by dividing the frequency of theoscillations by the number of oscillations that are supposed to occur ina scan line and comparing the quotient frequency to the occurrence ofhorizontal sync pulses to develop an error signal that is lowpassfiltered to generate the AFPC voltage. For example, the frequency of theVCO oscillations can be divided by using a digital counter to count eachof their crossings of their average-value axis in a prescribeddirection. A folding carrier switching between +1 and −1 to provide a 5MHz square wave can be generated in accordance with one of the leastsignificant bits of the digital counter. Another digital counter can bearranged to count vertical sync pulses separated from the luminancesignal, thereby generating a modulo-two frame count for controlling theframe-to-frame alternation of the folding carrier phase. A scan linecounter can be arranged to count horizontal sync pulses separated fromthe luminance signal, and the least significant bit of the scan linecount can be used for controlling the line-to-line alternation of thefolding carrier phase.

FIG. 5 is a more detailed block diagram of a portion of the FIG. 2circuitry. In FIG. 5, an input terminal 205 connects from the outputterminal of the ADC 102 of FIG. 2 to receive digitized composite videosignal V. Input terminal 205, is coupled to respective input terminalsof a vertical highpass filter (VHPF) 202, a temporal highpass filter(THPF) 204, a horizontal bandpass filter (HBPF) 206 and to respectiveminuend input terminals of subtractors 208 and 210. An output terminalof the VHPF 202 is coupled to an input terminal of a horizontal highpassfilter (HHPF) 212, which may have a break frequency of 1.7 MHz, forexample,. An output terminal of HHPF 212 is coupled to a subtrahendinput terminal of subtractor 208. An output terminal of subtractor 208is coupled to an input terminal of a horizontal lowpass filter (HLPF)209. An output terminal of HLPF 209 is coupled to a first data inputterminal of a soft switch 214. An output terminal of soft switch 214 iscoupled to an output terminal 215. Output terminal 215 is coupled to theinput terminal of the deemphasis an folding circuitry 108 of FIG. 2.

An output terminal of THPF 204 is coupled to an input terminal of ahorizontal highpass filter (HHPF) 216 and to a minuend input terminal ofa subtractor 218. An output terminal of HHPF 216 is coupled torespective subtrahend input terminals of subtractors 210 and 218. Anoutput terminal of subtractor 210 is coupled to a second data inputterminal of soft switch 214.

An output terminal of subtractor 218 is coupled to an input of a signalmagnitude detector (rectifier) 220. An output terminal of magnitudedetector 220 is coupled to an input terminal of a signal spreader 222.An output terminal of signal spreader 222 is coupled to an outputterminal 225 and to a control input terminal of soft switch 214. Outputterminal 225 is coupled to the motion signal input terminal ofchrominance/motion signal combining circuit 11 6 of FIG. 2.

An output terminal of HBPF 206 is coupled to an input terminal of ananti-crosstalk processor 224 for processing the chrominance componentprior to combining with the motion signal.

An output terminal of anti-crosstalk processor 224 is coupled to aninput terminal of a chrominance signal modulator 226 forming a part ofthe chrominance/motion signal combining circuit 11 6 of FIG. 2 thatconverts the 3.58 MHz color subcarrier sidebands to a color-under signalcentered at 629 kHz, as will be described in detail further on in thisspecification.

The horizontal HPFs 212 and 216 may be standard digital highpassfilters, each having a break frequency at around 2 MHz. A 15-taphorizontal highpass filter is preferred, yielding a responsecharacteristic which is 6 dB at 1.75 MHz. In regard to the diagonalspatial prefiltering performed in the encoder prior to folding and alsothe diagonal spatial post-filtering after unfolding (as will bedescribed later), the diagonal filters in the encoder and decoder havematched filter characteristics. In the diagonal filtering process, theinput signal is vertically highpass-filtered by VHPF 202; then thevertically high-passed part of the signal is horizontallyhighpass-filtered with HHPF 212; then the resultant signal is subtractedfrom the input signal in the subtractor 208 to provide a diagonallylowpass filtered output signal; and in turn the diagonally lowpassfiltered output signal is horizontally lowpass-filtered by HLPF 209,having a break frequency at around 3.3 MHz, to produce thespatially-filtered luminance signal Y_(S).

In operation, the horizontally and vertically highpass filtered signalHVHP produced by the cascaded VHPF 202 and HHPF 212 contains all thechrominance information present in the composite video signal V inaddition to all the spatial detail information. This signal HV_(HP) issubtracted from the composite video signal V by differencing insubtractor 208 to produce a diagonally lowpass-filtered signal HV_(LP)containing only the luminance information. The output signal HV_(LP)from subtractor 208 is applied to HLPF 209. HLPF 209 removes thehorizontal frequency spectrum components above its break frequency (3.3MHz, for example) from HV_(LP) to avoid aliasing noise in the spatiallyreconstructed luminance signal during playback processing, therebyproviding at the output of HLPF 209 a spatially-filtered luminancesignal Y_(S).

The spatially-filtered luminance signal Y_(S) produced by HLPF 209therefore contains only luminance information, but has reduced diagonalresolution. Temporally and horizontally highpass filtered signalHT_(HP), produced by the cascaded THPF 204 and HHPF 216, also containsall the chrominance information present in the composite video signal V,in addition to most of the temporal detail information. This signalHT_(HP) is subtracted from the composite video signal V by differencingin subtractor 210 to produce a temporally-filtered luminance signal. Thetemporally-filtered luminance signal Y_(T) produced by subtractor 210therefore contains only luminance information at full spatialresolution, but has reduced temporal resolution.

The temporally highpass filtered signal T_(HP) from THPF 204 containsmotion information at horizontal low frequencies and chrominanceinformation at high horizontal frequencies. Thus, the output signalHT_(HP) from HHPF 216 is subtracted by differencing in subtractor 218from the temporally highpass filtered signal T_(hp) to derive ahorizontally lowpass-filtered, temporally highpass-filtered signalH_(LP)T_(HP) which is a bipolar motion-representative signal. The signalH_(LP)T_(HP) varies in magnitude as a function of both the magnitude ofthe motion in the image (that is, the greater the degree of motion inthe image, the greater the signal magnitude) and the contrast betweenthe moving and still portions of the image. This signal H_(LP)T_(HP) hasgreatest magnitude at the edges of an object having high contrast withrespect to the background against which it is moving. Where thebackground and the moving object are close in intensity, themotion-representative signal H_(LP)T_(HP) has a low magnitude. Inaddition, quickly moving objects with soft edges also produce a lowmagnitude motion signal. Finally, even with quickly moving, highcontrast objects, the motion-representative signal H_(LP)T_(HP) isusually only strong within several pixels of the moving object's edge.

In order to minimize the effect of these variations in the-filteredmotion-representative signal H_(LP)T_(HP), magnitude detector 220detects the magnitude of the motion-representative signal H_(LP)T_(HP)from the subtractor 218 and produces a single-bit signal indicatingeither the presence or absence of motion for that pixel. A knownmagnitude detector 220 may include a multiplexer having a control inputterminal responsive to a sign bit of the applied motion- representativesignal H_(LP)T_(HP). The motion-representative signal H_(LP)T_(HP) wouldbe coupled to a first input terminal of the multiplexer and an inputterminal of an arithmetic negator circuit. An output terminal of thearithmetic negator circuit would be coupled to a second input terminalof the multiplexer. The output terminal of the multiplexer produces themagnitude (absolute value) of the motion- representative signalH_(LP)T_(HP). If the sign bit is a logic “0”, indicating, for example,that the motion-representative signal value is positive, then themultiplexer couples the first input terminal, carrying themotion-representative signal H_(LP)T_(HP) to the output terminal. If thesign bit is a logic “1”, indicating that the motion-representativesignal value is negative, then the multiplexer couples the second inputterminal, carrying the arithmetic negative of the motion-representativesignal H_(LP)T_(HP) (which would be a positive valued signal) from thenegator to the output terminal.

This magnitude signal is then supplied to a known comparator circuit.The comparator circuit compares the magnitude signal to a predeterminedthreshold value. If the magnitude signal exceeds the threshold value,then the comparator circuit produces an output signal which is a logic“1” signal. If the magnitude signal is less than the threshold value,then the comparator circuit produces an output signal which is a logic“0” signal. The output of this comparator is a single bitmotion-representative signal which is a logic “1” in the presence ofmotion, and a logic “0” otherwise.

This single-bit motion-representative signal is spread vertically andhorizontally by signal spreader 222 to generate the spread motion signalM. Optionally, the signal may be spread temporally, vertically andhorizontally by signal spreader 222. Apparatus for spreading such asingle bit motion-representative-signal is described in detail in U.S.Pat. No. 5,083,203 issued Jan. 21, 1992 to J. W. Ko et alii and entitledCONTROL SIGNAL SPREADER, and assigned to Samsung Electronics Co., Ltd.,pursuant to the obligations of the inventors to so assign theirinventions at the time those inventions were made. The spread motionsignal M produced by signal spreader 222 is a multi-bit digital signalwhose value gradually decreases from a maximum value in moving areas (asindicated by the single-bit bi-level signal having a logic “1” value) toa minimum (zero) value in the still region area around the moving areain the vertical and horizontal directions (and optionally, temporally).This spread motion signal M is used in the adaptive luma separator 104for adaptively processing the video signal V as described below. Themotion signal M is also compatibly encoded so as to be recordable andreproducible, to be recovered and utilized by a decoder as will bedescribed in detail later.

As described above, in the absence of image motion, the luminance signalY is preferably the temporally-filtered luminance signal Y_(T), but inthe presence of image motion, the luminance signal Y is preferably thespatially-filtered luminance signal Y_(S). Soft switch 214 willcontinuously vary the proportion of the two input signals Y_(T) andY_(S) which can be coupled to the luminance signal Y output terminal 215in response to the value of the motion signal M. If the value of themotion signal M is zero, or nearly zero, indicating no motion or a lowlevel of motion, then the soft switch produces an output signal Y whichis composed completely of the temporally-filtered luminance input signalY_(T). If the value of the motion signal M is at a maximum, or nearlymaximum, indicating a high level of motion, then the soft switch 214produces an output signal Y which is composed completely of thespatially-filtered luminance signal Y_(S). At intermediate values of themotion signal M, the output signal contains some proportion of each ofthe input signals Y_(T) and Y_(S). The operation of soft switch 214 willbe described in more detail below.

The NTSC chrominance component is extracted from the composite videosignal V in a known manner using the horizontal, BPF 206. The separatedchrominance component signal (modulated on the 3.58 MHz NTSC chromasubcarrier) from horizontal BPF 206 is processed to avoid crosstalk byanti-crosstalk processor 224, and then supplied as chrominance signal Cto the input of the chroma modulator 226 of chrominance/motion signalcombining circuit 116, to be down-converted in frequency to a 629 kHzcolor-under signal for VHS format recording by chrominance signalmodulator 226. This can be done in a known manner—e. g., by heterodyning3.58 MHz NTSC chroma signal against a 4.21 MHz four-phase carrier andselecting only the lower resultant sidebands, to provide the color-underchrominance component signal amplitude modulated on a 629 kHz carrier.Before its down-conversion by the modulator 226, the chrominancecomponent signal is processed by the anti-crosstalk element 224 toreduce adjacent line crosstalk with the motion signal M in the compositechrominance-plus-motion signal (C+M). Anti-crosstalk element 224 may be,for example, a vertical highpass filter (VHPF), which may be implementedas a three-tap line-comb lowpass filter. Optionally, a verticalfiltering of the composite video signal V may precede the horizontalbandpass filtering by horizontal BPF 206 at the chroma separation stage.

FIG. 6 shows a particular construction of the FIG. 5 circuitry in whichthe comb filters VHPF 202 and THPF 204, both responsive to the compositevideo signal V are implemented so they can share delay lines. In FIG. 6,elements which are the same as those in FIG. 5 have the same referencenumber designation and are not described in further detail below.

In FIG. 6, THPF 204 is shown as including a subtractor 230 having aminuend input terminal connected to the input terminal 205 for receivingthe digitized composite video signal V and having a subtrahend inputterminal for receiving the signal V as delayed by one frame. The signalV is delayed by one frame by the cascade connection of a first 1H delayline 232, a second 1H delay line 234 and a 1F-2H delay line 236, alsoincluded in THPF 204. The 1H delay lines 232 and 234 each provide arespective delay equal to the duration of one horizontal scan line; andthe 1F-2H delay line 236 provides a respective delay shorter than theduration of one frame scan by the duration of two horizontal scan lines.The difference output of the subtractor 230 supplies the response T_(HP)of THPF 204.

VHPF 202 also incorporates the 1H delay lines 232 and 234, sharing themwith THPF 204, as well as including an adder 238 and a subtractor 240not included in THPF 204. The digitized composite video signal V asreceived at terminal 205 is halved by a bit place shift for applicationas a first summand input to the adder 238; and the signal V is delayedby the duration of two horizontal scan lines in the 1H delay lines 232and 234, then halved by a bit place shift for application as a secondsummand input to the adder 238. The sum output signal of the adder 238is halved by a bit place shift for application as a subtrahend inputsignal of the subtractor 240. The digitized composite video signal Vreceived at terminal 205 and delayed by the duration of one horizontalscan line in the 1H delay line 232 is halved by a bit place shift forapplication as a minuend input signal of the subtractor 240. Thedifference output signal from the subtractor 240 is supplied to theinput terminal of HHPF 212 as the response of VHPF 202, when VHPF 202 isimplemented per FIG. 6.

The separated chrominance and luminance detail information response atthe output terminal of HHPF 212 is supplied as the subtrahend inputsignal for the subtractor 208 in FIG. 6, just as in FIG. 5. Thedigitized composite video signal V received at terminal 205 and delayedby the duration of one horizontal scan line in the 1H delay line 232 isfurther delayed (by means not shown in FIG. 6) to compensate for thedelays through the subtractor 240 and HHPF 212, and after being sofurther delayed is supplied as a minuend input signal for the subtractor208. The difference output signal from the subtractor 208 is adiagonally lowpass filtered output signal horizontally lowpass-filteredby HLPF 209 to produce the spatially-filtered luminance signal Y_(S).

In FIG. 6 the output terminal of subtractor 218 is followed by thecascade connection of the magnitude detector 220, a horizontal spreader242 and a vertical spreader 244. The combination of horizontal spreader242 and vertical spreader 244 forms the motion signal spreader 222 ofFIG. 5 and operates as described above. The remainder of the FIG. 6block diagram is the same as illustrated in FIG. 5 and described above.It will be understood that FIG. 6 does not purport to show timingaccuracy, or timing matching. That is, FIG. 6 does not show all thedelay lines which would be utilized for equalizing the delays along therespective signal paths to maintain pixel correlation. A person ofordinary skill in the signal processing art would understand the needfor providing correction for timing mismatching, and would also haveknowledge of various ways in which such correction could be implemented,and it is therefore not necessary to describe such in detail here.

FIG. 7 is a more detailed block diagram of the soft switch 214illustrated in FIG. 5. Soft switch 214 is utilized for motion-adaptiveprocessing of the spatially-filtered luminance signal Y_(S) and thetemporally-filtered luminance signal Y_(T). In FIG. 7, a first signalinput terminal 405 of soft switch 214 is coupled to the output terminalof subtractor 210 of FIG. 5 for receiving the temporally-filteredluminance signal Y_(T) therefrom. Input terminal 405 is coupled to afirst input terminal of a multiplier 404. An output terminal ofmultiplier 404 is coupled to a first input terminal of an adder 412. Anoutput terminal of adder 412 is coupled to an output terminal 435.Output terminal 435 is coupled to the deemphasis and folding circuitry108 of FIG. 2.

A second signal input terminal 415 of soft switch 214 is coupled to anoutput terminal of the HLPF 209 of FIG. 5 for receiving thespatially-filtered luminance signal Y_(S) therefrom.

Input terminal 415 is coupled to a first input terminal of a multiplier408. An output terminal of multiplier 408 is coupled to a second inputterminal of adder 412. A control input terminal 425 of soft switch 214is coupled to the spread motion signal (M) output terminal of signalspreader 222 of FIG. 5. Input terminal 425 is coupled to an inputterminal of a read-only memory (ROM) 410 storing a look-up table. Aportion of the readout from the read-only memory 410 provides a firstscaling factor K to a second input terminal of multiplier 404, andanother portion of the readout from the read-only memory 410 provides asecond scaling factor (1−K) to a second input terminal of multiplier408.

In operation, the spread motion signal M from input terminal 425 isapplied to the address input of a read-only memory 410 storing a look-uptable of the two scaling factors, K and (1−K) which are related to thevalue of the control signal M. The first scaling factor K is theproportion of the temporally-filtered luminance signal Y_(T) whichshould be in the luminance output signal Y. The second scaling factor(1−K) is the proportion of the spatially-filtered luminance signal Y_(S)which should be in the motion-adapted luminance output signal Y. The sumof K and (1−K) is one. The multiplier 404 scales the temporally-filteredluminance signal Y_(T) by the scaling factor K, and multiplier 408scales the spatially-filtered luminance signal Y_(S) by the scalingfactor (1−K). Adder 412 sums the two scaled signals output by themultipliers 404 and 408 to produce the motion-adaptivelyspatio-temporally filtered separated luminance signal Y.

The motion adaptive spatio-temporal luminance processing function K(M)is selected such that when M is equal to zero or nearly zero(corresponding to a low level of motion in the luminance component), Kis equal to one (all temporally-filtered luminance) and (I−K) is equalto zero (no spatially-filtered luminance), and when M is at maximum ornearly maximum (corresponding to a high level of motion in the luminancecomponent), K is equal to zero (no temporally-filtered luminance) and(1−K) is equal to one (all spatially-filtered luminance). The functionK(M) is continuous and may be linear or non-linear. As the value of themotion signal M gradually changes from zero to maximum, the proportionof the temporally-filtered luminance signal Y_(T) in the luminanceoutput signal Y gradually decreases and the proportion of thespatially-filtered luminance signal Y_(S) in the luminance signal outputY gradually increases, and vice versa.

In a modification of the FIG. 5 circuitry, the second scaling factor(1−K) instead of being stored in ROM could be generated by subtractingfrom unity the first scaling factor K as supplied from ROM addressed bythe spread motion signal M. Or, alternatively, the first scaling factorK instead of being stored in ROM could be generated by subtracting fromunity the second scaling factor (1−K) as supplied from ROM addressed bythe spread motion signal M.

As described above, during recording it is possible to derive amotion-representative signal during the processing of the inputcomposite video signal to separate its luminance signal component.So-called “false motion” may be introduced into thismotion-representative signal by the chrominance signal (i. e.,chrominance information which aliases as motion), but this erroneousdetection of motion can be largely eliminated by temporallylowpass-filtering the spatially lowpass-filtered composite signal asshown in FIG. 5 (or, alternatively, by vertically and horizontallylowpass filtering the temporally highpass-filtered signal). Because theNTSC chrominance component sidebands do not extend down below 2 MHz, thehorizontal lowpass filtering ensures that chrominance components whichmight give rise to false motion are removed from themotion-representative signal during the motion derivation process.

Also as described above, the luminance high frequencies are folded intothe low-frequency luminance signal spectrum by modulating them on afolding carrier which is placed in a Fukinuki hole, similar to themanner in which the NTSC chrominance subcarrier is placed in thecomposite NTSC video signal. However, there are no restrictions on howfar down in frequency the lower sidebands of the folded luminance highfrequencies can extend. In fact, diagonal detail in the high-bandluminance signal, when folded into the luminance low-band frequencies,can extend all the way down to spatial DC. Because the folding carrieris alternated on a frame-to-frame basis (to maximize the temporaldistance from DC) these diagonal details may be erroneously detected asmotion, and no degree of spatial filtering can suppress this type of“false motion”. Thus, to properly remove the folding byproducts from theunfolded luminance signal on playback, it is necessary to supply aseparate channel for passing the motion-representative signal, derivedduring the encoding side processing and utilized for motion-adaptivelyfiltering the separated luminance component signal, to the decoder sidefor utilization in motion adaptively prefiltering the unfolded luminancesignal.

Providing a separate channel for supplying the motion representativesignal to the playback circuitry facilitates the maintenance ofcorrespondence of the motion-adaptive luminance reconstruction processin the playback circuitry to the motion-adaptive processing used in theluminance signal separator of the record circuitry. For example, supposethe luminance signal separator in the record circuitry uses temporalprocessing in some region of the image to derive the luminance signalwith extended resolution in the diagonal directions. It would beundesirable to use spatial processing on the playback side toreconstruct in the same region of the image, since the extendedresolution in the diagonal directions would be discarded, makingfruitless the effort made during recording to preserve such resolution.

Further, the chrominance/luminance signal separation process performedon the composite NTSC signal in the encoder, no matter how well done,can introduce some artifacts into the image (e. g., chrominance aliasingas luminance and luminance aliasing as chrominance. The full-bandwidthluminance signal reconstruction process performed in the decoder canalso introduce artifacts into the image. If the second process isindependent of the first process, then the artifacts introduced by theupstream process have artifacts introduced upon them by the downstreamprocess, intensifying them. Artifact intensification can be greatlyreduced if the downstream processing can be made to faithfully follow,that is, to parallel, the upstream processing.

In an implementation compatible with the VHS format, a separate channelin the transmission or recording medium for the motion representativesignal is provided for by encoding the motion signal into vacantquadrants of the VHS format color-under carrier on the record side. Onthe playback side, the encoded motion signal must be separated from thereproduced color-under carrier component, in order that the recoveredmotion signal may be utilized for motion-adaptive filtering of theunfolded luminance signal.

FIG. 8 shows the chrominance/motion combining circuit 116 of FIG. 2according to a VHS format-compatible embodiment in more detail. Thespread motion signal M from output terminal 225 is input to a modulator610 and modulated on a 250 kHz four-phase carrier to generate amodulated motion signal component having a horizontal frequency of 250kHz and with its phase shifting forward or backward 90° per line inalternate fields in complementary fashion (phase complement) to that ofthe 629 VHS kHz color-under carrier C, so that in those fields (tracks)where the color-under carrier C occupies the first and third quadrants,the motion signal M occupies the second and fourth quadrants, while inopposite fields (tracks) the color-under carrier C and motion signal Mreverse quadrants. The down-converted color-under carrier component Cfrom chroma modulator 226 in FIG. 5 and the modulated motion signalcomponent M from modulator 610 are input to respective signal inputs ofan adder 620 to be combined into the resultant chrominance-plus-motionsignal (C+M). One skilled in the art that the chrominance signal C maybe appropriately processed for color burst emphasis—by way of example,using burst emphasis or gating circuitry 235 to boost the amplitude ofcolor burst portions of the chrominance signal C prior to that signalbeing supplied to adder 620. The chrominance-plus-motion signal (C+M)supplied from the adder 620 thus contains the chrominance information Cas well as the spread motion signal information M modulated on afour-phase carrier but occupying complementary quadrants of the carrier,advancing and retarding 90° in phase and alternating between even andodd quadrants in alternate fields. The chrominance-plus-motion signal(C+M) from the adder 620 is then filtered by a horizontal lowpass filter(HLPF) 630 having a break frequency around 1.2 to 1.3 MHz. The responseof the filter 630 supplied to DAC 118 to be converted to an analogsignal (C+M)R included as an under signal in the frequency-multiplexedsignal supplied to the VCR 40 for recording.

The choice of the 250 kHz carrier frequency for the motion signal ismade in order best to reduce the visibility of interference duringslowest playback of an encoded recording on a conventional VHS formatplayback device. However, it is also possible to modulate the motioninformation on a 629 kHz carrier like the chroma, so long as thequadrants occupied by the respective signals are complementary asdescribed above. This procedure is satisfactory except at the slowestplayback speeds. The generation of under signals encoding chrominancesignal C and motion signal M is described in further detail in theabove-referenced U.S. patent application Ser. No. 059,765 filed May 11,1993.

FIG. 9 is a more detailed block diagram of the signal separator 50, theunfolder 60 and the reemphasis circuitry 70 following the VCP 45 in FIG.1. In FIG. 9, an input terminal 800 receives the frequency-multiplexsignal reproduced by the VCP 45 of FIG. 1 when playing back from thevideo tape cassette.

Input terminal 800 is coupled to the input terminal of a horizontalhighpass filter 802 that separates the frequency-modulated luminancecarrier from the reproduced frequency-multiplex signal for applicationto FM detector circuitry 803, which circuitry typically comprises alimiter amplifier and an edge-counting FM detector. The folded luminancesignal Y_(PB) recovered by the FM detector circuitry 803 is supplied tothe input terminal of an analog-to-digital converter (ADC) 804 . Anoutput terminal of ADC 804 is coupled to an input terminal of ahorizontal lowpass filter (HLPF) 805 having a passband cutting off ataround 2.5 MHz or 3 MHz, depending on the bandwidth of the foldedluminance signal Y_(PB). An advantage of filtering the playbackluminance signal digitally is that the group delay characteristics ofthe digital HLPF 805 may be made flat, which is difficult to achieve inan analog implementation. The output terminal of HLPF 805 is coupled toan input terminal of a time base corrector (TBC) 806. An output terminalof TBC 806 is coupled to a data input terminal of an unfolding circuit808. An output terminal of unfolding circuit 808 is coupled to aluminance signal input terminal of a spatio-temporal post-filter 820. Anoutput terminal of post-filter 820 is coupled to an input terminal of anadaptive reemphasis circuit 822. An output terminal of the adaptivereemphasis circuit 822 is coupled to a luminance signal input terminalof a composite video signal generator 810 for supplying it a (digital)reproduced luminance signal Y* (where “*” indicates a playback signalrepresenting the same signal as previously recorded on the cassette).The composite video signal generator 810 of a known type combines adigital chrominance signal C* with the luminance signal Y* to form astandard (digital or analog) composite video signal. Composite videosignal generator 810 typically includes a DAC or DACs for converting theapplied digital luminance and chrominance component signals Y* and C* toanalog signals.

An output terminal of composite video signal generator 810 is coupled toan output terminal 815. Output terminal 815 is coupled to utilizationcircuitry (not shown) which, for example, may be a television receiverfor reproducing the images which were previously recorded on thecassette or a Y-C output jack.

Alternatively, the recovered luminance and chrominance signals Y* andC*, which are in digital form, may be output directly in digital formfor utilization in further processing, or may be converted to analogform by respective DACs and output directly as analog Y and C signalsfor further utilization.

Input terminal 800 is coupled to the input terminal of a horizontallowpass filter 801 that separates the under signal component encoding(C+M) from that reproduced frequency-multiplex signal. An outputterminal of the lowpass filter 801 connects to an input terminal of ananalog-to digital converter (ADC) 814, to supply for digitization theunder signal component encoding (C+M) separated from the reproducedfrequency-multiplex signal. An output terminal of ADC 814 is coupled toan input terminal of a time base corrector (TBC) 816. An output terminalof TBC 816 is coupled to an input terminal of a chrominance/motionsignal separator 818. The chrominance/motion signal separator 818includes an upconverter for generating a digitized chrominance signal C*from the color-under signal. A chrominance signal output terminal of thechrominance/motion signal separator 818 is coupled to a chrominanceinput terminal of the composite video signal generator 810 for supplyingit the digitized chrominance signal C*. A motion signal output terminalof the chrominance/motion signal separator 818 is coupled to a controlinput terminal of the spatio-temporal prefilter 820.

In operation, the elements 802, 803, 804, 805, 806, 808, 820 and 822 inFIG. 9 operate to extract the full-bandwidth luminance signal from thereduced bandwidth luminance signal previously recorded on the cassette.ADC 804 produces a sampled multi-bit digital signal representing theplayback folded luminance signal. The TBC 806 operates to correct anytiming inaccuracies which are introduced by jitter in the tape mechanismor any other source of timing inaccuracy, and produces the recoveredfolded luminance signal Y_(F)*.

After time base correction by TBC 806, the folded luminance signalY_(F)* Is applied to one input of the unfolding circuit 808. Thelow-band luminance component tends to be in-phase from frame to frame,and the folded luminance high-band component tends to reverse phase fromframe to frame. The unfolding circuit 808 unfolds (i. e., re-shifts) theluminance high-band frequencies which were previously folded into theluminance low-band frequency spectrum and combines the low-band andhigh-band luminance signals to output the full-bandwidth unfoldedluminance signal Y_(UF). This full-bandwidth unfolded luminance signalY_(UF) is supplied to the spatio-temporal post-filter 820 where theunfolded full-bandwidth luminance signal Y_(UF) is motion-adaptivelyspatio-temporally filtered to provide the deemphasized luminance signalY_(D)* having the high-frequency luminance components still deemphasizeddue to the record side deemphasis processing. This unfolded deemphasizedluminance signal Y_(D)* is supplied to the adaptive reemphasis circuit822 where the deemphasized, high-frequency components are adaptivelyreemphasized to restore them to their original amplitude to provide therecovered full-bandwidth luminance signal Y* with proper amplituderelationship.

The recovered full-bandwidth luminance signal Y* is supplied to theluminance signal input terminal of the composite video signal generator810. Composite video signal generator 810 operates in a known manner tocombine the luminance signal Y* and chrominance signal C* to form astandard (digital or analog) composite video signal. This signal may beused by any equipment which utilizes such a signal, for example, atelevision receiver or display monitor.

The elements 801, 814, 816 and 818 in FIG. 9 operate to extract thechrominance-plus-motion (C+M) signal previously recorded on thecassette. The ADC 814 produces a sampled multi-bit digital signalrepresenting the chrominance-plus-motion signal and the TBC 816 operatesto correct any timing inaccuracies in this signal, and produces therecovered chrominance-plus-motion signal (C+M)*. A chrominance/motionseparator 818 processes the recovered chrominance-plus-motion signal(C+M)* to produce a recovered motion signal M*, which is supplied to thecontrol input terminal of the spatio-temporal post-filter 820, and toup-convert the color-under signal to produce the chrominance signal C*,which is supplied to the chrominance signal input terminal of thecomposite video signal generator 810.

When recorded, the chrominance signal and the luminance signals were inphase synchronism. However, they are passed through two separateindependent paths in the record circuitry (illustrated in FIG. 1) andare frequency-division-multiplexed on the cassette. This separateprocessing may introduce phase inaccuracies between the two signalswhich are not compensated for in the two separate TBCs 806 and 816.Apparatus for restoring the proper phase relationship between thechrominance and luminance signals is described in detail in U.S. Pat.No. 5,083,197 issued Jan. 21, 1992 to J. W. Ko et alii, entitledAPPARATUS FOR RESTORING THE CORRECT PHASE RELATION OF THE CHROMA ANDLUMINANCE SIGNALS PASSED THROUGH SEPARATE PATHS, and assigned to SamsungElectronics Co., Ltd., pursuant to the obligations of the inventors toso assign their inventions at the time those inventions were made.

FIG. 10 is a block diagram of a portion of the luminance recoverysection illustrated in the upper half of FIG. 9, showing in more detailthe unfolding circuit 808 and the spatio-temporal post-filter 820. Aftertime base correction by TBC 806, the folded luminance signal Y_(F)* isapplied to one input of unfolding circuit 808, which may be implementedby a modulator 902 to which is also supplied an unfolding carrier havinga frequency f_(U). The folded luminance signal is unfolded by direct or“straight” sub-Nyquist sampling Y_(F)* (as contrasted to the “offset”technique employed during folding) around the unfolding frequency(selected to be 5 MHz, for example, in accordance with the criteriadescribed above in the description of the folding modulator 512 of FIG.35) by the modulator 902, to provide the unfolded luminance signalY_(UF).

Unfolding modulator 902 may be constructed in a known manner using afour quadrant multiplier, and is preferably a +1, 0 type modulatoroperating to insert zero values replacing odd or even samples dependingon the unfolding phase, driven by a clock signal at one-half thesampling frequency, which in this example is a sampling frequency of 10MHz.

The unfolded (i. e., remodulated) luminance signal Y_(UF) is thenapplied to the input terminal of spatio-temporal post-filter 820 forremoval of byproducts of the unfolding process prior to reemphasis ofthe unfolded high-frequency luminance component. The post-filter 820includes a temporal lowpass filter (TLPF) 904 configured as a frame-comblowpass filter (which may be identical in structure and operation toTHPF 204, HHPF 216 and subtractor 210 in FIGS. 5 and 6) which providesframe averaging and removes components all the way down to spatial DCfrom the unfolded luminance signal Y_(UF) for providingtemporally-filtered unfolded luminance signal Y_(T)*. Temporal filter904 is arranged in parallel with a spatial filter (SPF) 906 (which maybe identical in structure and operation to VHPF 202, HHPF 212,subtractor 208 and HLPF 209 in FIGS. 5 and 6) acting as a diagonallowpass filter for providing spatially-filtered unfolded luminancesignal Y_(S)*.

A soft switch 914 having its data inputs connected to the Y_(T)* andY_(S)* outputs of TMF 904 and SPF 906, respectively, varies its dataoutput proportionally between the temporally filtered and spatiallyfiltered unfolded luminance signals applied at its data inputs from TMF904 and SPF 906, under control of motion-adaptive scaling factor signalsK* and (1−K)* which are applied to control input terminals of softswitch 914 from a look-up table (LUT) stored in a read-only memory 910.The ROM 910 generates scaling factor signals K* and (1−K)* for each LUTtable entry stored therein, as individually addressed in accordance withthe recovered motion signal M* supplied to the input address terminalsof ROM 910 from chrominance/motion separation circuit 818, forperforming motion-adaptive post-filtering of the unfolded luminancesignal Y_(UF) prior to the reemphasis stage processing. The outputterminal of soft switch 914 is coupled to an output terminal ofpost-filter 820 at which is provided the spatio-temporally post-filteredunfolded deemphasized luminance signal Y_(D)* which is in turn coupledto the luminance signal input terminal of adaptive reemphasis circuit822.

As noted, SPF 906 may correspond in structure and operation to theluminance spatial filter section of the adaptive luma separation circuit104 formed by VHPF 202, HHPF 212, subtractor 208 and HLPF 209 shown inFIG. 5. SPF 906 provides a diagonal lowpass filter response forspatially processing the unfolded signal to remove unfoldingartifacts—i. e., remodulation byproducts and residual unfolding carrierwhich may be present during image motion and which manifest strongly inthe diagonal. This correspondence permits utilization of the same filterin encoding and decoding. Soft switch 914 controls the proportion of thetemporally-filtered and spatially-filtered unfolded full-bandwidthluminance signals Y_(T)* and Y_(S)* to be included in thespatio-temporally post-filtered unfolded deemphasized luminance signalY_(D)* in response to the recovered motion control signal M*. When thelevel of image motion is zero or nearly zero, the output of the softswitch 914 consists entirely of temporally-filtered unfolded luminancesignal Y_(T)* from TMF 904, and does not contain any ofspatially-filtered unfolded luminance signal Y_(S)*. As the magnitude ofmotion in the image gradually increases, the proportion of thetemporally-filtered luminance signal Y_(T)* input from the TMF 904 inthe output of the soft switch 914 correspondingly decreases and theproportion of the spatially-filtered luminance signal Y_(S)* input fromSPF 906 correspondingly increases. In the presence of relatively highlevels of motion, the output from the soft switch 914 will consistentirely of the spatially-filtered signal Y_(S)* from SPF 906.

Referring now to FIG. 11, there is shown a more detailed block diagramof the soft switch 914 of the post-filter 820 in FIG. 9 or 10. Softswitch 914 is constructed in identical fashion to soft switch 214 shownin FIG. 7. In FIG. 11, input terminals 1005 and 1015 fortemporally-filtered unfolded luminance signal Y_(T)* andspatially-filtered unfolded luminance signal Y_(S)* respectivelycorrespond to input terminals 405 and 415 in FIG. 7. Multipliers 1004and 1008 correspond to multipliers 404 and 408, respectively, in FIG. 7.Likewise, ROM 910 in

FIG. 11 having motion signal M* input terminal 1025 corresponds to ROM410 in FIG. 5 having motion signal M input terminal 425.

Similarly, adder 1012 in FIG. 11 corresponds to adder 412 in FIG. 7,summing the outputs of modulators 1004 and 1008 to provide thereby themotion adaptively spatio-temporally filtered unfolded deemphasizedluminance signal Y_(D)* at output terminal 1012 . Generation of thescaling factor signals K* and (1−K)* by ROM 910 in accordance with theapplied recovered motion signal M* is performed in the same manner aswas previously described above with regard to the operation of softswitch 214 in FIG. 7 for generating scaling factors K and (1−K) inaccordance with motion signal M, and will not be described in furtherdetail here.

Advantageously, the ROMs 410 and 910 respectively used during recordingand during playback may be one and the same. Furthermore, because thecombining circuitry 30 and signal separator 50 will typically not beoperated simultaneously, for advantages of convenience and economy theprefilter 820 in the playback electronics may share many common elementssuch as filter blocks and the soft switch with the luminance separationcircuit 104 in the recording electronics.

The separation of the composite chrominance-plus-motion signal (C+M)* inchrominance/motion separation circuit 818 of FIG. 9 as implemented forapplication to use with the conventional VHS format will now bedescribed in more detail with reference to FIGS. 12 and 13. In FIG. 12the separation process is performed directly on the (C+M)* signal priorto up-conversion, while it is still at the color-under frequency. Asshown in FIG. 12, the chrominance-plus-motion signal (C+M)* from TBC 816is applied to the input of a quadrant selective filter QSF 1202 and alsoto the minuend input of a subtractor 1204. QSF 1202 selects between oddand even quadrants of the input (C+M)* signal and may be implemented bya diagonal filter with delay of 2H (i. e., twice the duration of ahorizontal scan line) and having four unique coefficients and astructure as shown in FIG. 13. The horizontal frequency response orselectivity of QSF 1202 is centered on 629 kHz. The bandwidth of thepass region is approximately 1 MHz, providing 500 kHz bandwidth for eachsideband for maintaining good chroma response. Due to the fixed natureof QSF 1202, it will be appreciated that for those reproduced tracks (i.e., fields) from the encoded recorded videotape in which the chrominancecomponent of the signal (C+M)* is in a quadrant of QSF 1202 with thepositive peaks (pass bands), the filter will pass only the chrominancecomponent, while in the next track (field) only the motion signalcomponent will be passed. Accordingly, QSF passes the motion componentor the chroma component, depending on whether the track being decoded isodd or even.

For each track (field/channel) reproduced by the VCP 45, QSF 1202 willpass either the C* or the M* component of the reproduced composite(C+M)* signal to its output, depending on which component is located inthe filter's pass region during that track. The output of QSF 1202 iscoupled to the subtrahend input of subtractor 1204 in order to obtainthe opposite signal component, M* or C*, by differencing the filteroutput against the composite (C+M)* signal applied at the minuend inputof subtractor 1204. The output of QSF 1202 (i. e., M* or C*) and theoutput of subtractor 1204 (i. e., C* or M*) are coupled to respectiveinputs of a multiplexer (MUX) 1206 having a pair of inputs and a pair ofoutputs which are switched at the field rate of 60 Hz under control of asignal (e. g., field pulse or channel-1/channel-2 switching signal)which may be generated in known manner by a conventional head switchingcircuit (not shown) of the VCP 45, or by conventional channel switchingcircuitry associated with playback preamplifier circuits. One output ofMUX 1206 therefore will continuously provide only the separatedchrominance component C* for each track/channel reproduced, while theother output of MUX 1206 will continuously provide only the separatedmotion signal component M*.

The separated motion signal M* is coupled to the input of a full-waverectifier 1208. As described further on with regard to FIG. 14 thefull-wave rectifier 1208 may comprise a synchronous detector for 250kHz. The output of the rectifier 1208 is coupled to the input of ahorizontal lowpass filter (HLPF) 1210 which may be implemented with 15taps at about 500 kHz. The recovered spread motion signal M* output fromHLPF 1210 is supplied to the address input of ROM 910 of thespatio-temporal post-filter 820 for controlling the motion-adaptivefiltering of the unfolded luminance signal Y_(UF) as previouslydescribed.

The separated chrominance signal component C* supplied from MUX 1206 maybe converted to an analog signal by a respective DAC and then processedby a conventional VHS chroma recovery circuit in known manner to obtainthe 3.58 MHz NTSC chroma component. This may in fact be preferable fromthe viewpoint of chroma phase control processing during picture-searchand still modes, which is more complex if performed digitally. However,the separated chrominance signal component C* output from MUX 1206 maybe also be digitally processed by a digital implementation of aconventional VHS chroma recovery circuit employing a modulator 1220 (e.g., a multiplier) supplied with a 4.21 MHz four-phase carrier, wherebythe 629 kHz chroma component carrier is frequency up-converted to 3.58MHz by heterodyning, then passed through a 3.58 MHz bandpass filter(BPF) 1222 to pass the 3.58 MHz chroma. The chrominance component may befiltered further for removing residual up-conversion carrier andmodulation byproducts, and processed for burst deemphasis, if desired.The recovered digital 3.58 MHz chrominance sub-carrier component signalmay then be converted to an analog NTSC chroma component signal by arespective DAC, supplied to composite video signal generator 810 (whichmay include a DAC), or utilized in further processing.

FIG. 14 shows apparatus that can be used in the record electronics of aVCR for generating the folding carrier f_(F). The same apparatus isincluded in the playback electronics of a VCR or VCP for generating anunfolding carrier suitable for unfolding luma highs from the Fukinukiareas. The unfolding carrier, which is the same as the folding carrier,is a harmonic of an even multiple of both the line and the frame scanrates, which harmonic reverses phase from scan line to scan line withineach frame and from frame to frame.

The generation of the unfolding carrier begins with a line-lockedvoltage-controlled oscillator (VCO) 300, which oscillates at 640 timesscan line frequency (i. e., at 10.07 MHz). Crossings of theiraverage-value axis by those oscillations, which crossings are in aprescribed direction, are detected by an average-axis crossing detector302 to generate pulses at 10.07 MHz rate that are used to time certainsampling procedures. These pulses are also supplied to a pixel counter304 for counting in a frequency-dividing procedure. (The word “pixel” isshort for “picture element”.) A decoder 306 generates a logic “1” whenthe pixel counter 304 count reaches the value indicating the counter 304has counted 640 pixels since last being reset. This logic “1” resets thepixel counter 304 to initial count on the next 10.07 MHz rate pulsesupplied to its count input, so the logic “1” condition exists as apulse of about one sample duration. This pulse is supplied from thedecoder 306 to a scan line counter 308 and to an AFPC detector 310.

The AFPC detector 310 compares this pulse (or a portion thereof, such asits trailing edge) with separated horizontal synchronizing pulsessupplied from a horizontal sync separator 312 to develop an errorvoltage, which is lowpass filtered by an AFPC loop filter 314. The AFPCloop filter 314 responds to the error voltage to supply automaticfrequency and phase control (AFPC) voltage for controlling the VCO 300,closing the AFPC loop. This AFPC loop “line-locks” the 10.07 MHzoscillations of the VCO 300 to the horizontal sync pulses occurring atthe 15,734 kHz scan line rate.

The scan line counter 308 is reset to initial count by the output signalfrom a two-input AND gate 322 going from logic “1” to logic “0”. Theleading edges of separated vertical synchronizing pulses supplied from avertical sync separator 318 are detected by a leading edge detector 320to generate a logic “1” applied as count input to a field counter 330.The output signal from the leading edge detector 320 is also applied asone input signal to the AND gate 322. A decoder 324 decodes the last 160pixels at the end of scan lines to generate a logic “1” at the time whenseparated vertical sync pulses supplied to the AND gate 322, if theybegin, must be those occurring at the beginning of a frame rather thanmidway through. The AND gate 322 accordingly generates a logic “1” pulseat the conclusion of each frame. This logic “1” resets the pixel counter308 to initial count on the next 15,734 kHz rate pulse supplied to itscount input from the decoder 306.

The least significant bit of the pixel counter 304 is applied to oneinput of an exclusive-OR gate 326. The other input of the XOR gate 326receives a sign bit that alternates between logic “1” and logic “0”respectively on successive scan lines. FIG. 14 shows this sign bit beingsupplied from the output of another exclusive-OR gate 328 receiving at afirst input thereof a modulo-two frame count bit from a field counter330 for controlling the frame-to-frame alternation of the foldingcarrier phase and receiving at a second input thereof the leastsignificant bit of the scan line count from the scan line counter 308for controlling the line-to-line alternation of the folding carrierphase, which scan line counter is assumed to be of the above-describedtype reset once per frame. Upon reception of the sign bit supplied fromthe exclusive-OR gate 328, the XOR gate 326 generates a two's complementf_(F) folding carrier +1, −1 of 5 MHz frequency in accordance with theleast significant bit supplied from the pixel counter 304 with its phasereverses at scan line rate, and each reversal being at a respectiveinstant between successive scan lines.

The field counter 330 is assumed to be a two-stage counter countingfields on a modulo-four basis, with the field count being supplied bothto a four-phase 4.21 MHz carrier generator and to a four-phase 250 kHzcarrier generator. The four-phase 4.21 MHz carrier generator generatesthe carrier used for up-converting the color-under signal to reproducethe chroma sidebands of the 3.58 MHz color subcarrier. The four-phase250 kHz carrier generator generates the carrier used for detecting the Msignal. The field count stored in the field counter 330 during playbackmust correspond to the field count during recording. The logic “1” theAND gate 322 generates at the conclusion of each frame resets thecounter stage supplying the less significant modulo-two-field-count bitof the modulo-four field counter 330 to “1” when the field counter 330next receives at its count input a logic “1” to logic “0” transitionfrom the leading edge detector 320. Error in the more significantmodulo-two-frame-count bit of the modulo-four field counter 330 iscorrected by a trigger pulse being applied from an AND gate 332 to thecounter stage supplying that more significant bit.

The fact of the more significant modulo-two-frame-count bit of themodulo-four field counter 330 being in error can be determined by thesign bit of the signal M* as synchronously detected in accordance withits four-phase 250 kHz carrier. The 250 kHz carrier is the sixteenthharmonic of horizontal scan line frequency f_(H). Carrier extractioncircuitry 334 responds to two appropriately selected bits of the pixelcount from the counter 304 to generate a repeating 0, +1, 0, −1 sequenceof samples descriptive of one phasing of 16f_(H), which repeatingsequence a phase shifter 336 shifts by one sample position eachsuccessive scan line responsive to the two least significant bits of thescan line count from the line counter 308 and the modulo-four fieldcount from the field counter 330. The four-phase repeating 0, +1, 0, −1sequence of samples from the phase shifter 336 is supplied to asynchronous detector 338 as the four-phase 250 kHz carrier against whichthe signal M* from the multiplexer 1206 is synchronously detectedthereby to rectify it. A horizontal lowpass filter 340 rejects the imageof the M* baseband signal and smoothes it against high-frequency noisethat can alter its sign bit.

The sign bit of the filter 340 response to the synchronously detected M*signal is supplied to one of the two inputs of the AND gate 332, theother input of which is supplied recurring triggering pulses (e. g.,from the horizontal sync separator, as shown in FIG. 14). The sign bitof the synchronously detected M* signal will invariably be a “0” if thefield count in the field counter 330 is properly synchronized with thefield count during recording. This will inhibit the AND gate 332 fromresponding to the recurring triggering pulses supplied to its otherinput to apply triggering to the counter stage supplying the moresignificant modulo-two-frame-count bit of the modulo-four field counter330. The sign bit of the synchronously detected M* signal will be a “1”if the field count in the field counter 330 is not properly synchronizedwith the field count during recording or will become a “1” within afield or so. The sign bit of the synchronously detected M* signal being“1” will enable the AND gate 332 to apply the next occurring triggeringpulse supplied to its other input to the counter stage supplying themore significant modulo-two-frame-count bit of the modulo-four fieldcounter 330, toggling the output condition of that stage to correct itsstate.

In the FIG. 14 apparatus included in the playback electronics of a VCRor VCP for generating an unfolding carrier, the horizontal syncseparator 312 and the vertical sync separator 318 respectively separatehorizontal and vertical synchronizing pulses the post-filtered unfoldedluminance signal Y_(D)* from the output of soft switch 914 shown inFIGS. 9 and 10. The signal Y_(D)* applied to the horizontal syncseparator 312 may be delayed by some portion of a scan line respectiveto the signal Y_(D)* applied to the vertical sync separator 318 to avoidtiming problems.

Apparatus included in the record electronics of a VCR for generating afolding carrier is generally similar to the FIG. 14 apparatus forgenerating an unfolding carrier, except as follows.

The horizontal sync separator and the vertical sync separatorrespectively separate horizontal and vertical synchronizing pulses fromthe signal Y or the signal Y_(D). Further, there is no need to correctthe more significant modulo-two-frame-count bit of the modulo-four fieldcounter, so elements corresponding to elements 332 through 340 can bedispensed with. Quite obviously, in a single-deck VCR the apparatus forgenerating a folding carrier and the apparatus for generating anunfolding carrier may be constructed so as to share many elements. Theapparatus for generating a folding carrier and the apparatus forgenerating an unfolding carrier can take a variety of forms, includinganalog circuit forms; many design variations from that specificallydescribed will readily occur to those skilled in the art of designingdigital circuits or video signal processing circuitry.

Next, the deemphasis and folding processing in accordance with an aspectof the invention will be described. As described above, in priorluminance signal folding systems, the luminance high frequencies werefolded back as aliases into the luminance low-frequency spectralbandwidth at their original amplitude (or, boosted to a higheramplitude, if pre-emphasized per the teaching of Howson and Bell). Ifsuch folded luminance signals are recorded and then played back on aconventional narrow bandwidth VCR which has no provision for removingthe folded high luminance frequency aliases, high objectionableartifacts will be present in reproduced images, preventing such a videorecording from being backward compatible for conventional playbackapparatus reproduction. By appropriately deemphasizing the foldedhigh-frequency luminance components in the band-limiting foldingoperation during the encoder side processing, it is advantageously madepossible to reduce the artifacts in the video image displayed on playingback the band-limited folded luminance signal on the videotape with aconventional narrow bandwidth reproduction device to a level which isnot objectionable to the viewer, thereby providing desirable backwardcompatibility with existing playback apparatus.

FIG. 15 shows adaptive deemphasis and folding circuitry 108 of a type inwhich folding of the luminance signal frequency spectrum is done afterthe upper frequencies have been adaptively deemphasized respective tothe lower frequencies. Except for preferably being carried outdigitally, the spectrum folding itself is done in substantial conformitywith the second technique described by Howson and Bell.

Spatio-temporally processed luminance signal Y from the output terminalof the adaptive luminance separator 104 of FIG. 2 (i. e., from theterminal 215 of soft switch 214 in FIG. 5) is supplied to an inputterminal 440 of the adaptive deemphasis circuitry 460. The deemphasiscircuitry 460 supplies deemphasized luminance signal Y_(D) from anoutput terminal thereof to an adder 450 as a summand input signalthereto. The deemphasis circuitry 460 also supplies the deemphasizedluminance signal Y_(D) to a modulator 470 as a modulating signal inputthereto, for modulating the amplitude of a folding carrier of frequency2f_(F) supplied to another input of the modulator 470. The modulator 470is preferably a four-quadrant multiplier in which the product outputsignal from which the folding carrier of frequency 2f_(F) is suppressed.When the folding carrier comprises a string of repeating +1, −1 samplesin each horizontal scan line the construction of the modulator 470becomes very simple, essentially consisting of a multiplexer choosingbetween the deemphasized luminance signal Y_(D) supplied thereto as amodulating signal input and −Y_(D) as derived therein from Y_(D),alternatively selecting Y_(D) and then −Y_(D).

The product output signal from the modulator 470 comprises a loweramplitude-modulation sideband, which is a reversed-spectrum signalextending down from the 2f_(F) folding carrier frequency to DC, and anupper amplitude-modulation sideband, which is a non-reversed-spectrumsignal extending up from the 2f_(F) folding carrier frequency to a4f_(F) upper frequency limit. This product output signal is supplied tothe adder 450 as another summand input signal thereto. The sum outputsignal of the adder 450 is supplied to the input terminal of ahorizontal lowpass filter 480, which has a passband extending at leastto the 2.5 MHz folding frequency f_(F). Preferably, for improvingplayback in accordance with reasons similar to those discussed furtheron in the application with reference to FIGS. 37 and 38, the pass bandof the filter 480 extends somewhat past the 2.5 MHz folding frequencyf_(F) to 3 MHz or so. The lowpass response of the filter 480 appears ata terminal 90 as the signal Y_(F). As described previously in regard toFIG. 2, the signal Y_(F) is supplied to the DAC 110 to be converted toan analog signal Y_(R), which modulates the frequency of the controlledoscillator 120.

FIG. 16 shows one type of adaptive deemphasis circuit with controlsignal generator, which adaptive deemphasis circuit is suitable for useas the adaptive deemphasis circuit 460 in FIG. and embodies an aspect ofthe invention. This type of adaptive deemphasis circuit filtersluminance signal Y received at its input terminal 501 from the adaptiveluma separator 104 of FIG. 2, to reduce the high frequencies of thatsignal respective to its low frequencies, and supplies the resultingresponse at its output terminal 520 to the adaptive deemphasis andfolding circuitry 108 of FIG. 2. More particularly, the luminance signalY from the soft switch 214 is supplied to the terminal 501. A horizontallowpass filter (HLPF) 502 responds to only the low-frequency (e. g., DCto 2.5 MHz) components of this signal Y to supply a low-band lumaresponse Y_(L). A horizontal highpass filter (HHPF) 504 responds to onlythe high-frequency (e. g., 2.5 to 5 MHz) components of this signal Y tosupply a high-band luma response Y_(H). The luma high frequencies aresupplied from the horizontal highpass filter 504 to the multiplier 508as an input multiplicand signal, to be mutiplied by a control signalgenerated by a control signal generator 510 in response to the energycontent in the high-band luma response Y_(H) HHPF 504 supplies to thegenerator 510. An adder 506 is used to combine the low-band lumaresponse Y_(L) from HLPF 502 with the deemphasized high-band lumaresponse Y_(HD) supplied as product output signal by the multiplier 508.

FIG. 17 shows an input full-bandwidth signal, and FIG. 18 shows the samesignal after folding of the high-frequency band by sub-Nyquist samplingto reduce it to one-half the bandwidth of the original, the foldedhigh-band signal component being indicated by the broken lines. In asimple implementation of a deemphasis circuit for band-limiting duringrecording, in accordance with an aspect of the present invention,multiplier 508 and control signal generator 510 may be replaced by asimple attenuator providing a fixed amount of attenuation or amplitudereduction to high-frequency luminance signal component Y_(H), SO thatthe high-frequency luminance amplitude level in the folded luminancesignal Y_(F) is maintained below a level at which objectionableartifacts would become noticeable in the displayed narrow bandwidthvideo image on reproduction by a narrow band playback device. FIG. 19shows the folded signal of FIG. 18 after it has undergone a fixeddeemphasis of the amplitude of the folded high-frequency components byapproximately one-half.

However, while providing against objectionable artifacts (i. e., dotcrawl) in the displayed narrow band image which could result from highamplitude folded high-frequency components in the folded narrow bandluminance signal, this fixed form of deemphasis can undesirably degradethe signal-to-noise ratio, S/N, on the playback side because portions ofthe high-band luminance signal which are at a relatively low amplitude,such as broad flat areas with little or no contrast change, may also bereduced in amplitude by the fixed deemphasis process, degrading the S/Nratio of these portions of the recorded luminance signal duringreproduction.

That is, because folding the high-band luminance at full amplitude willcause a severe disturbance in the reproduced images on conventionalplayback, as occurs in the case of the system described by Faroudja, itis therefore desirable, for preventing the folded high-frequencyluminance components from manifesting as objectionable artifacts in thedisplayed image when playing back an encoded recording on a conventionalnarrow bandwidth VCR, to reduce the amplitude (i. e., modulation level)of the high-band luminance component interleaved with the low-bandluminance component. Reducing the modulation level of the folded highsby one-half can provide improved backward compatibility, but at the costof increased noise in the displayed wide bandwidth improved image. Thisnoise increase occurs when the attenuated highs are boosted in theplayback side decoding for restoring them to their original level. Thisnoise is most noticeable in broad, low level, flat areas of the image.

In accordance with an aspect of the present invention, the deemphasizingof the folded high-band luminance component during the record sideencoding and the accompanying reemphasis processing for restoring theunfolded deemphasized high-band luminance component to its originalamplitude during playback side decoding are preferably performedadaptively. i.e., the level of the folded high-band luminance componentis deemphasized adaptively during the encoding processing, andreemphasized adaptively during the decoding processing. Such adaptivedeemphasis of the high-band luminance component during the foldingprocess can significantly improve image noise performance during decodedplayback of the encoded recording by a playback system in accordancewith an aspect of the invention, while also providing enhanced backwardcompatibility to the encoded recording.

The adaptive deemphasis processing provides folding of the high-bandluma Y_(H) at a full level when the high-band luminance component is ata very low amplitude, and folding of the high-band luma Y_(H) at areduced level when the high-band luminance is at a high amplitude. Whenthe high-band luminance is adaptively deemphasized in this manner priorto or during the folding process, the reemphasis operation duringplayback decoding only increases the noise level during high-frequencyhigh-amplitude transitions where it is not noticeable on viewing thereproduced image.

FIG. 20 shows the folded high-band component of FIG. 18 after adaptivedeemphasis in accordance with an aspect of the invention, from which itmay be seen that the level of deemphasis of the folded high-band lumaY_(H) is varied in comparison to the fixed deemphasis in FIG. 19. FIG.21 corresponds to FIG. 20 but shows additionally the effect of a noisecoring operation performed in adaptively deemphasizing the foldedhigh-band luma Y_(H).

FIGS. 22, 23 and 24 illustrate ways in which the FIG. 16 band-splittingfilter comprising the HLPF 502 and the HHPF 504 may be constructed indigital form so as to save hardware, which ways require that the −6 dBfrequencies of the HLPF 502 and the HHPF 504 be the same. The FIG. 22,23 and 24 band-splitting filters are tapped-delay-line filters.

In FIG. 22 tapped delay line 5001 with an odd number of taps receives atits input the Y signal at the terminal 501. The taps are considered asbeing consecutively numbered beginning at one on the input end andincrementing by one each successive tap, which successive tap exhibits aone-sample delay from the tap before. The odd-numbered taps supply theirsignals to a weighted summation circuit 5002 for weighting those sampleswith positive weights that are alternate samples of both the symmetricalkernel of the HLPF 502 and the symmetrical kernel of the HHPF 504. Theeven-numbered taps supply their signals to a weighted summation circuit5003 for weighting those samples with positive weights that arealternate samples of the symmetrical kernel of the HLPF 502 and that arethe negatives of alternate samples the symmetrical kernel of the HHPF504. An adder 5004 receives as summands the weighted sums from theweighted summation circuits 5002 and 5003, adding them to generate theHLPF 502 response. A subtractor 5005 receives as minuend the weightedsum from the weighted summation circuit 5002, receives as subtrahend theweighted sum from the weighted summation circuit 5003, and supplies theHHPF 504 response as the difference between these weighted sums.

In FIG. 23 the HLPF 502 is a linear-phase finite-impulse-response (FIR)filter which comprises a tapped delay line 5021 with an odd number oftaps spaced at one-sample intervals and a weighted summation circuit5022 for weighting those samples with positive weights to provide asymmetrical kernel. The tapped delay line 5021 receives at its input theY signal at the terminal 501. The weighted summation circuit 5022supplies the HLPF 502 response for application to the adder 506 in theFIG. 16 adaptive deemphasis circuit. The central tap of the tapped delayline 5021 supplies Y signal delayed the same amount with respect to theY signal at the terminal 501 as the HLPF 502 response is; and asubtractor 503 subtracts the HLPF 502 response from this delayed Ysignal to obtain horizontal highpass frequency response. In the FIG. 16adaptive deemphasis circuit the difference output signal from thesubtractor 503 is applied to the multiplier 508 as its multiplicandsignal and is also applied to the control signal generator 510 as itsinput signal.

In FIG. 24 the HHPF 504 is a linear-phase finite-impulse-response (FIR)filter which comprises a tapped delay line 5041 with an odd number oftaps spaced at one-sample intervals and a weighted summation circuit5042 for weighting those samples with alternately negative and positiveweights to provide a symmetrical kernel. The tapped delay line 5041receives at its input the Y signal at the terminal 501. In the FIG. 16adaptive deemphasis circuit, the weighted summation circuit 5042supplies the HHPF 504 response applied in the FIG. 16 adaptivedeemphasis circuit to the multiplier 508 as its multiplicand signal andto the control signal generator 510 as its input signal. The central tapof the tapped delay line 5041 supplies Y signal delayed the same amountwith respect to the Y signal at the terminal 501 as the HHPF 504response is; and a subtractor 505 subtracts the HHPF 504 response fromthis delayed Y signal to obtain horizontal highpass frequency responsefor application to the adder 506 in the FIG. 16 adaptive deemphasiscircuit.

FIG. 25 is a block diagram of a specific form the FIG. 16 adaptivedeemphasis circuitry may take when used as the adaptive deemphasiscircuit 10 and control signal generator 80 in the video signalrecording/playback system illustrated in FIG. 1. In FIG. 25, inputterminal 501 corresponds to input terminal 5 of FIG. 1.

Input terminal 501 connects to the input terminals of HLPF 502 and HHPF504. An output terminal of HLPF 502 connects to an input terminal of theadder 506 and to a first input terminal of the multiplier 508. An outputterminal of the adder 506 connects to the output terminal 520. Outputterminal 520 connects to the input terminal of the folding circuit 20 ofFIG. 1.

FIG. 25 shows the control signal generator 510 as including a rectifier512, a corer 513, a low-pass filter (LPF) 514, and an inverse functioncircuit 515. An output terminal of HHPF 504 connects to an inputterminal of the rectifier 512. An output terminal of the rectifier 512connects to an input terminal of the corer 513. An output terminal ofthe corer 513 connects to an input terminal of the LPF 514. An outputterminal of the LPF 514 produces a control signal G, and connects to aninput terminal of a (1−G) function circuit 515. An output terminal ofthe (1−G) function circuit 515 connects to a second input terminal ofthe multiplier 508. An output terminal of multiplier 508 connects to asecond input terminal of the adder 506.

The operation of the adaptive deemphasis circuit 10 and the controlsignal generator 80 of FIG. 1, as exemplified in FIG. 25, may be betterunderstood by referring to the waveform diagrams illustrated in FIG. 26.The HLPF 502 and the HHPF 504 respond separately to the low-frequencyand high-frequency portions of the luminance signal. The output signalfrom HLPF 502 contains only the low-frequency portion of the luminancesignal referred to as “low-band luma”. The output signal from HHPF 504contains only the high-frequency portion of the luminance signalreferred to as “high-band luma”.

The low-band luma is supplied to the adder 506 for combination withvariably attenuated high-band luma to form the deemphasized luminancesignal Y_(D) supplied to the terminal 520. The high-band luma maycontain luminance information representing vertical edges.

FIG. 26A illustrates two examples of vertical edges. On the left handside, a large amplitude vertical edge is illustrated and on the righthand side, a small vertical edge is illustrated. FIG. 26B illustratesthe signal at the output terminal of the HHPF 504. As FIG. 26B shows,this high-pass filter response contains no low-frequency components norany direct term of its own and exhibits excursions in bothpositive-going and negative-going directions with regard to an averagevalue associated with the absence of signal.

The absolute value of this high-pass filter response is obtained byrectifying the response, using the rectifier 512. FIG. 26C illustratesthe signal at the output terminal of rectifier 512.

In digital circuitry the rectifier 512 takes the form of anabsolute-value circuit. If the digital samples descriptive of luminancehigh frequencies use two's complement arithmetic, one known way thatrectification can be done digitally is by exclusive-ORing the sign bitwith each of the less significant bits, to generate respective bits of anumber to which a ZERO sign bit is affixed and unity is then added. Theaddition generates the absolute-value response to the input signal intwo's complement arithmetic.

The corer 513 receives the rectified luminance high frequencies from therectifier 512 and operates as a type of threshold circuit sometimesreferred to as a “baseline clipper”, thereby to eliminate the effect oflow-amplitude edges on the deemphasis function. If the value of theinput signal from the rectifier 512 is less than the threshold value,then a zero-valued signal is produced by the corer 513. If the value ofthe input signal from the rectifier 512 is greater than the thresholdvalue, then the value of the corer 513 output signal is the value of theinput signal less the threshold value. FIG. 26D illustrates the signalat the output terminal of corer 513. Referring to FIG. 26C, the dottedline illustrates the threshold value. In FIG. 26D, only the portion ofrectified signal illustrated in FIG. 26C which exceeds the thresholdvalue passes through the corer 513, producing the signal illustrated inFIG. 26D.

This type of coring is well known in analog circuitry. This type ofcoring as realized in digital circuitry replaces samples below athreshold value by zeroes and reduces the amplitude of samples not belowthe threshold value by the threshold value. This type of coring differsfrom the type of coring often encountered in digital processing inwhich, when generating the cored signal, input signal samples below athreshold value are replaced by zeroes and samples not below thethreshold value are reproduced without change. When the corer 513 is tobe of a type suitable for inclusion in digital circuitry, it can berealized using a digital subtractor and a multiplexer. The digitalsubtractor is connected for subtracting the threshold value from therectifier 512 output signal. The multiplexer is controlled by the signbit from the digital subtractor, generating the cored signal byselecting positive values of the difference output signal from thedigital subtractor for inclusion in the cored signal and selectingzeroes for inclusion in the cored signal rather than negative values ofthe difference output signal from the digital subtractor.

The output signal from corer 513 is supplied to a horizontal low-passfilter 514 to produce control signal G illustrated in FIG. 26E. The HLPF514 operates to spread corer 513 output signal so that the controlsignal G changes gradually in the vicinity of the vertical edge. Thecontrol signal G is scaled to vary between zero, when the level of theluminance signal is small, and a value G_(MAX) nearly one-half, when thelevel of the high-frequency portion is large. The control signal G isthen subtracted from the value one in the (1−G) function circuit 515,producing the signal illustrated in FIG. 26F. The (1−G) function circuit515 may be constructed in known manner of analog or digital arithmeticelements, or, in one type of digital implementation may be provided by alook-up table stored in read-only memory. This signal varies betweenone, when the level of the high-frequency portion of the luminancesignal is small, to a value (1−G_(MAX)) somewhat more than one-half whenthe level of the high-frequency portion is large. This signal issupplied to one input terminal of the multiplier 508. The multiplier 508scales the level of the high-frequency portion of the luminance signalby multiplying the high-frequency portion of the luminance signal by the(1−G) signal. This scaled high-frequency portion is then added to thelow-frequency portion of the luminance signal to form the deemphasizedluminance signal.

When the level of the high-frequency portion of the luminance signal ishigh, the scaling factor approaches one-half and the level of thehigh-frequency portion is nearly halved. When the level of thehigh-frequency portion of the luminance signal is low, the scalingfactor is nearly one and the level of the high-frequency portion ispassed through unattenuated. When the high-frequency portion of theluminance signal is at intermediate levels, the scaling factor isintermediate between one-half and one and the level of thehigh-frequency portion in the deemphasized luminance signal is at acorresponding intermediate value.

The determination of the value G_(MAX) will now be described in moredetail. When the peak amplitude of the separated high-frequency portionof the luma signal is denoted as Y_(H), the threshold level of the corer513 is denoted as T_(D), and g denotes the gain or scaling factorassociated with the cascade connection of the corer 513 and the low-passfilter 514, the product signal P the multiplier 508 supplies to theadder 506 is expressed by the equation (1), following.

P=Y_(H)[1−9(Y_(H)−T_(D))]

=(1+gT_(D)) Y_(H)−g Y_(H) ²  (1)

T_(D) is a fraction of one, and Y_(H) ranges from T_(D) to one. G_(MAX)is the value of the term g(Y_(H)−T_(D)) in the function P when Y_(H) hasits maximum value of one.

G_(MAX)=g(1−T_(D))  (2)

The linearity of the function P respective to Y_(H) is its slope,obtained by taking its derivative P′ respective to Y_(H).

P′=(1+gT_(D))−2g Y_(H)  (3)

It is desirable that P be monotonically increasing with Y_(H) across itsrange from zero to one, so each value of P is associated with only onevalue of Y_(H); this implies that P′ must never be negative. Since P′decreases with increase in Y_(H), the value of P′ when Y_(H) has itsmaximum value of one must be zero or more than zero.

(1+gT_(D))−2g>0  (4)

This expression can be rearranged to derive an expression on theboundary conditions for g in order that the function P monotonicallyincrease for Y_(H) increasing across its range from zero to one.

(1+gT_(D))>2g  (5)

1>2g−gT_(D)  (6)

1>(2−T_(D))g  (7)

(2−T_(D))⁻¹>g  (8)

g<(2−T_(D))⁻¹ (9)

That is, if the threshold T_(D) were zero, g could be no larger thanone-half. So, if the threshold T_(D) were zero, G_(MAX) would have thevalue g which would be one-half. This limit condition is approachedquite closely when the threshold T_(D) is close to zero.

For values of T_(D) larger than zero, G_(MAX) has to be reduced tosomewhat less than one-half. Combining equation (2) with the expression(9) of inequality, the following expression (10) of inequality isobtained.

G_(MAX)<(1−T_(D))(2−T_(D))⁻¹  (10)

For T_(D)=({fraction (1/16)}), G_(MAX)=({fraction (15/31)}) and thehigh-band luma signal Y_(H) can be reduced to as little as ({fraction(16/31)}) its original level without affecting the monotonicity of thecompression function P. For T_(D)=(⅛), G_(MAX)=({fraction (7/15)}) andthe high-band luma signal Y_(H) can be reduced to as little as({fraction (8/15)}) its original level without affecting themonotonicity of the compression function P. For T_(D)=(¼),G_(MAX)=({fraction (3/7)}) and the high-band luma signal Y_(H) can bereduced to as little as ({fraction (4/7)}) its original level withoutaffecting the monotonicity of the compression function P. If the (1−G)function circuit 515 is provided by a look-up table stored in read-onlymemory, the transfer function stored in the look-up table can bemodified so it is no longer a linear transfer function, reducing thecompression function P for values of Y_(H) that are intermediate betweenT_(D) and one, so the function P can be kept monotonic though thehigh-band luma signal Y_(H) is reduced to one half or even less of itsoriginal level. Since the linearity of the reemphasis of the high-bandluma signal Y_(H) by the reemphasis circuits shown in FIGS. 27, 28 and29 depends directly on the linearity of the compression function P, suchmodification of the transfer function stored in the look-up table alsoimproves the fidelity of the reemphasized luminance signal to theluminance signal received for recording before being processed by theFIG. circuitry to deemphasize its high-frequency portion Y_(H).

The above-referenced U.S. patent application Ser. No. 008,813 usesadaptive deemphasis circuitry of the type shown in FIG. 25 in a videotape recording and playback system characterized by the signal G beingencoded in a second under signal, This second under signal is includedwith the frequency-modulated luminance carrier and color-under signal ina frequency multiplex signal recorded on the video tape in accordancewith modified VHS recording procedures.

FIG. 27 shows an embodiment of the FIG. 1 reemphasis circuit 70. Theoutput terminal of the FIG. 1 unfolding circuit 60 connects to an inputterminal 1100 shown in FIG. 27. Input terminal 1100 connects torespective input terminals of a horizontal lowpass filter (HLPF) 1102and of a high-pass filter (HHPF) 1104. An output terminal of horizontallowpass filter (HLPF) 1102 connects to a first input terminal of anadder 1106. An output terminal of the adder 1106 connects to outputterminal 15, which corresponds to the output terminal 15 of FIG. 1. Anoutput terminal of the HHPF 1104 connects to an input terminal of acorer 1113. An output terminal of corer 1113 connects to a first inputterminal of multiplier 1118. A source of a scaling factor is connectedto a second input terminal of multiplier 1118.

In operation, the HHPF 1104 and HLPF 1102 respectively respond only tothe low-frequency and high-frequency portions of the reproduced unfoldeddeemphasized luminance signal from each other. The response of the HPF1104 to the reproduced unfolded deemphasized luminance signal containsno low-frequency components nor any direct term of its own, exhibitsexcursions in both positive-going and negative-going directions withregard to an average value associated with the absence of signal andreferred to conventionally as “zero” value, and is subjected to a coringoperation in corer 1113.

Corer 1113 operates as a bidirectional threshold circuit, as opposed tocorer 513 (of FIG. 25) which operates only in the positive signaldirection. If the value of the input signal is closer to zero or averagevalue than a predetermined threshold value T_(R), then a zero-valuedsignal is produced. If the value of the input signal is farther fromzero or average value than the threshold value T_(R), then the value ofthe output signal is the value of the input signal less the thresholdvalue T_(R). FIG. 21G illustrates the signal at the input terminal ofcorer 1113 and the positive and negative thresholds as referred to anaverage value associated with the absence of signal and referred toconventionally as “zero” value. In FIG. 21G, the corer circuit respondsonly to the portion of input signal which exceeds the threshold value.

The coring operation by the corer 1113 removes low-amplitude noise inthe high-frequency portion of luminance resulting from the recording andplayback process. Such noise is particularly noticeable in areas of theimage where there is no detail. During recording the adaptive deemphasisdid not reduce the amplitude of the high-frequency portion of luminancein areas of the image where the level of the high-frequency portion wassmall. Therefore, in these areas the level of the high-frequency portionof luminance will tend to remain higher than the noise. In the FIG. 27circuit the threshold amplitude T_(R) is set to be about one-half thethreshold amplitude T_(D) used during recording. The corer 1113 willrespond, then, only to excursions of the separated high-frequencyportion of luminance signal that go below −T_(R) and that go above TR.The coring operation performed by the corer 1113 removes a higherfraction of the energy of the high-frequency portion of the deemphasizedluminance where it exceeds threshold amplitude T_(D) by a small amountthan where it exceeds threshold amplitude T_(D) by a greater amount;this phenomenon tends to restore in some degree the dynamic range in theluminance high-frequencies that was lost during deemphasis.

The corer 1113 may be constructed in analog circuitry by adjusting thelevel of the corer input signal supplied to a symmetrical clipper orlimiter and then differentially responding to the corer input signalbefore and after its being clipped or limited (e. g. with an operationalamplifier configured as an analog subtractor) to generate the coredresponse. Other analog coring circuits are known.

The type of coring performed by the corer 1113 differs from the type ofcoring usually encountered in digital processing in which, whengenerating the cored signal, input signal samples having amplitudessmaller than a threshold value are replaced by zeroes and samples havingamplitudes larger than the threshold value are reproduced withoutchange. Digital-signal coring in which samples having amplitudes largerthan the threshold value are reproduced without change, removessubstantially no energy from edge transients having amplitudes above thethreshold value, which is desirable in most simple noise-coringoperations. In the FIG. 27 reemphasis circuit, however, the goal ofrestoring dynamic range for the luminance high frequencies requirescoring of a type that removes a substantial fraction of the energy fromedge transients having amplitudes not much above the threshold valueT_(R), but not from edge transients having amplitudes the absolutevalues of which are way above the threshold value T_(R).

In digital circuitry the corer 1113 may be implemented by using alook-up table (LUT) stored in ROM or by using circuitry for performingsuitable arithmetic and logical operations. As an example of how thelatter course can be pursued, the corer 1113 can constructed as follows,using two's complement arithmetic, so as to remove a substantialfraction of the energy from edge transients having amplitudes not muchabove the threshold value, but so as not to remove a reduced fraction ofthe energy from edge transients further above the threshold value. In adigital subtractor a positive digital threshold value is subtracted fromthe corer 1113 input signal, and in a first digital adder the corer 1113input signal is added to the positive threshold value. Responsive to aZERO sign bit in the difference signal from the subtractor, a firstmultiplexer selects that difference signal as its output signal; andresponsive to a ONE sign bit in the difference signal from thesubtractor, the first multiplexer selects arithmetic zero as its outputsignal. Responsive to a ZERO sign bit in the sum signal from the firstadder, a second multiplexer selects arithmetic zero as its outputsignal; and responsive to a ONE sign bit in the sum signal from thefirst adder, the second multiplexer selects that sum signal as itsoutput signal.

A second digital adder adds the output signals of the digital subtractorand first digital adder to generate the cored response.

The cored high-frequency portion of the deemphasized luminance signal isscaled up by multiplier 1118 to compensate more or less for the energyof the high-frequency luminance lost in the coring by corer 1113. Thescaling factor is applied to the scaling factor input terminal ofmultiplier 1118. This scaling factor may be fixed at a predeterminedlevel between one and three. Preferably, the scaling factor is fixed attwo when the threshold levels in the corer 1113 are set to halve theenergy of the high-frequency portion of luminance recorded with a (1−G)scaling factor just less than one. This provides accurate scaling of thehigh-frequency portion of luminance recorded with a (1−G) scalingfactors of one-half and just less than one. This scaled high-frequencyportion from multiplier 1118 is recombined with the low-frequencyportion from subtractor 1102 in adder 1106. The output from adder 1106is the full-bandwidth reemphasized luminance signal.

In alternative embodiments of the FIG. 27 reemphasis circuitry, insteadof having fixed threshold values in the corer 1113 and a fixed scalingfactor in the multiplier 1118, provision may be made for theseparameters to be controlled by a user. A user threshold adjust inputterminal 1115 may be coupled to a threshold input terminal of corer1113, as illustrated in phantom in FIG. 27. A threshold adjust signal,supplied to the threshold adjust input terminal 1115, under the controlof a user, will adjust the threshold of corer 1113. If the threshold isadjusted too low, then noise will appear in areas of the image having nodetail; if the threshold is adjusted too high, then only the highestamplitude detail will be provided in the image, and the image may assumea smeared look. The user may adjust the threshold to produce the mostpleasing image.

Also, a user gain adjust input terminal 1120 may be coupled to thescaling factor input terminal of multiplier 1118, illustrated in phantomin FIG. 27. A gain adjust signal, supplied to the gain adjust inputterminal 1115, under the control of a user, will adjust the scalingfactor. As described above, preferably this scaling factor can beselected by a user from a range extending between one and three,inclusive. Ganged controls for the threshold adjust and gain adjustsignals can be used instead of individual user-adjusted controls.

In still further alternatives to the FIG. 27 reemphasis circuit, aplurality of corers with differing threshold levels and with differingscalings of their respective output signal levels as summed with thelow-frequency portion of the unfolded luminance signal supplied from theHLPF 1102 can be used instead of the single corer 1113 and scalingmultiplier 1108. This approach allows reemphasis to be exact at morethan two levels of high-frequency luma. This approach tends to be morefeasible in digital circuitry than in analog circuitry. However, itusually simpler to use a look-up table stored in read-only memory toobtain more sophisticated reemphasis characteristics, as will bedescribed further on in this specification.

The FIG. 28 reemphasis circuit replaces the single corer 1113 andscaling multiplier 1118 with: a corer 1123 and scaling multiplier 1128for passing luminance signal high frequencies that are aboveaccompanying noise, and with a corer 1133 and scaling multiplier 1138for augmenting luminance signal high frequencies that are above thethreshold level used in adaptive deemphasis. This augmentation expandsthe dynamic range of the high frequencies of the luminance signal at theoutput terminal 15 so as to restore the dynamic range to besubstantially the same as that of the luminance signal as originallyreceived for recording. The two-input adder 1106 is replaced by athree-input adder 1116 for combining the low-frequency luminance signalcomponent from the subtractor 1102 and the high-frequency luminancesignal components from the scaling multipliers 1128 and 1138 to form thefull-band luminance signal the adder 1106 supplies to the outputterminal 15. The coring levels of the corers 1123 and 1133 are shown asbeing adjustable responsive to user-adjusted signals applied to 1125 andto threshold adjust input terminals 1125 and 1115, respectively, and themultiplier 1128 receives a user-adjusted signal applied to gain adjustinput terminal 1120.

The threshold levels between which the corer 1123 provides coring areadjusted to closely bracket the zero-signal, or average, level of thecompressed-in-dynamic-range high frequencies of luminance signalseparated by the HHPF 1104. This is so noise unaccompanied by actualluminance signal high frequencies is suppressed in output signal fromthe corer 1123, while only a relatively small amount of energy isremoved from those high frequencies that are substantially above theaccompanying noise. A first scaling factor, which is little more thanone, is applied by the multiplier 1128 to the output signal from thecorer 1123 before the application of that signal to the adder 1116 as asummand input signal. Indeed, the output signal from the corer 1123 maybe directly supplied to the adder 1116 as a summand input signal withoutany scaling up in amplitude; and the multiplier 1128, dispensed with. Auser-determined noise threshold signal descriptive of the absolutevalues of the threshold levels between which the corer 1128 providescoring (as referred to zero-signal, or average, level of thecompressed-in-dynamic-range high frequencies of luminance signalseparated by the HHPF 1104) may be supplied to the corer 1128 forsetting those threshold levels between which the corer 1123 providescoring. Providing for user control of the first scaling factor appliedby the multiplier 1128 is not really necessary; where provided, suchuser control is preferably ganged together with the noise thresholdsignal adjustment control.

When the FIG. 28 reemphasis circuit is constructed using digitalcircuitry, the corer 1123 can be of the type usually encountered indigital processing. In this usual type of digital-signal corer, whengenerating the cored signal, input signal samples having amplitudessmaller than a threshold value are replaced by zeroes and samples havingamplitudes larger than the threshold value are reproduced withoutchange. This coring procedure removes less energy from samples havingamplitudes larger than the threshold value and further reduces need forthe scaling multiplier 1128.

The corer 1133, however, is used for restoring dynamic range for theluminance high frequencies and thus requires coring of a type thatremoves a substantial fraction of the energy from edge transients havingamplitudes not much above the threshold value, but not from edgetransients way above the threshold value. The threshold levels betweenwhich the corer 1133 provides coring can depart substantially from thezero-signal, or average, level of the compressed-in-dynamic-range highfrequencies of luminance signal separated by the HHPF 1104. Thesedepartures are large enough that the corer 1133 suppresses entirely theluminance signal high frequencies having amplitudes that are below thethreshold level used in adaptive deemphasis. That is, the thresholdlevel T_(R) for the corer 1133 is set at least as high as the thresholdlevel T_(D) for the corer 513 used in the FIG. 25 recording circuitry.Therefore, only from the corer 1123 does the adder 1116 receive responseto the luminance signal high frequencies that have amplitudes less thanthe threshold level T_(D) used in adaptive deemphasis. So, except fornoise coring by the corer 1123, these lower-level luminance signal highfrequencies pass through the system without compression or subsequentexpansion.

When the luminance signal high frequencies are of amplitude exceedingthe threshold level T_(D) used in adaptive deemphasis, the corer 1133supplies to the scaling multiplier 1138 response to the positive andnegative excursion peaks extending beyond the limits of its coringregion. A second scaling factor nominally one is applied to these peaksby the scaling multiplier 1138 before their application to the adder1116 as another summand input signal containing luminance signal highfrequencies for augmenting those in the summand input signal suppliedfrom the corer 1123 and scaling multiplier 1128. The higher-levelluminance signal high-frequency peaks, compressed to half their originallevel during recording are doubled during playback in an expansion thatrestores them substantially to their original level. Note that there isno boosting of high-frequency noise when reemphasizing luminance signalswith high-level high-frequency content when the threshold level T_(R)for the corer 1133 is set at the threshold level T_(D) for the corer 513used in the FIG. 25 recording circuitry.

One can set the threshold level T_(R) for the corer 1133 somewhat abovethe threshold level T_(D) for the corer 513 and can boost the corer 1133response somewhat before adding it back to full-spectrum luma in theadder 1116. This increases the error in high-frequency luma withamplitude just above the threshold level T_(R), but decreases the errorin high-frequency luma with higher amplitudes.

The FIG. 28 reemphasis circuitry can be modified to use additionalcoring at higher threshold levels, to detect higher-energy lumatransients for scaling and addition to the full-spectrum luma to furtherboost higher frequency transients.

Providing for user control of the second scaling factor applied by themultiplier 1128 is not really necessary; where provided, such usercontrol is preferably ganged together with a coring adjust control. Thescaling multiplier 1138 can be dispensed with, with output signal fromthe corer 1133 directly supplied to the adder 1126 as a summand inputsignal without any scaling in amplitude. User control via the terminal1125 of the limits of the coring region in input signal supplied to thecorer 1123 accommodates variations in video recording and playbackapparatus.

Self-adjusting reemphasis circuitry of the FIG. 28 type is possible. Theenvelope of the luminance signal high frequencies separated by the HHPF1104 can be detected, and the troughs of that envelope can be detectedto determine a noise coring adjustment signal for defining the coringregion of the input signal supplied to the corer 1123. The coringthresholds in the corer 513 of modified FIG. 25 adaptive deemphasiscircuitry and in the corer 1133 of modified FIG. 28 reemphasis circuitrycan be tracked by reference to a shared standard reference level. Suchshared standard reference level can be determined by scaling down fromthe change between synchronizing pulse tip and back porch levels in theluminance signal, for example, without further modification of VHSrecording standards. Or standard reference level pulses can beincorporated in a vertical blanking interval (VBI) scan line, if one iswilling to make further modification of VHS recording standards.

FIG. 29 shows reemphasis circuitry that does not core high-frequencynoise from luminance signals with low-level high-frequency content, butalso does not boost high-frequency noise when reemphasizing luminancesignals with high-level high-frequency content. HHPF 1104 responds tothe unfolded luminance signal received at input terminal 1100 toseparate the compressed-in-dynamic-range high-frequency portiontherefrom for application to a corer 1143, and the corer 1143 respondsonly to those luminance signal high frequencies that are smaller inamplitude than the threshold level used in adaptive deemphasis. An adder1126 adds the corer 1143 response as multiplied by the scalingmultiplier 1148 to the unfolded luminance signal received at the inputterminal 1100, thereby to generate a sum output signal supplied asreemphasized luminance signal from the output terminal corresponding tothe output terminal 15 of FIG. 1. The FIG. 28 reemphasis circuit, whenits noise threshold adjustment is set so the corer 1123 does not corethe luminance signal high-frequencies separated by the HHPF 1104,performs the same as the FIG. 29 reemphasis circuit.

The FIG. 29 reemphasis circuit may be modified to include a symmetricallimiter or slicer circuit that limits the response of the HHPF 1104 justabove noise level, possibly as adjusted by user or automatic control,which limited response is then subtractively combined with the luminancesignal supplied to the terminal 15.

This provides a way, alternative to that used in the FIG. 28 reemphasiscircuitry, for coring low-level high-frequency noise from the luminancesignal supplied to the terminal 15.

It is noted above that, if the (1−G) function circuit 515 is provided bya look-up table stored in read-only memory, the transfer function storedin the look-up table can be modified so it is no longer a lineartransfer function. This, it is pointed out, reduces the compressionfunction P for values of Y_(H) that are intermediate between T_(D) andone, so the function P can be kept monotonic though the luminance signalis reduced to one half or even less of its original level. The FIG. 25adaptive deemphasis circuit can be modified by reversing the order inwhich the corer 513 and the HLPF 514 are connected in cascade. Thisjuxtaposes the corer 513 and the (1−G) function circuit 515, so that aread-only memory can replace both of them. The G signal will no longerbe available to be used as a recorded control signal for use in anadaptive reemphasis circuit as described in U.S. patent application Ser.No. 008,813 filed Jan. 25, 1993. However, the response of therepositioned HLPF 514, or the output signal supplied from the ROMreplacing the corer 513 and the (1−G) function circuit 515, can be usedas a recorded control signal. The recorded control signal will then haveto be appropriately processed for use in the adaptive reemphasiscircuit, which processing can be done using a look-up table stored in

FIG. 30 shows the control signal generator 510 comprising cascadeconnection of a full-wave rectifier 512, a horizontal lowpass filter(HLPF) 514 and read-only memory (ROM) 516 storing the function r in alook-up table (LUT). The high-band luminance signal Y_(H) supplied tothe rectifier 512 is full-wave rectified and then applied to HLPF 514which has a break frequency of approximately 1 MHz. The output signalE_(H) from HLPF 514 provides an accurate representation of the averageenergy in the luminance high-band signal Y_(H) over a time perioddetermined by the time constant of HLPF 514. That is, the value ofsignal E_(H) represents the average “local” energy of Y_(H). In broadflat areas, the signal E_(H) will be at zero, whereas during sharpcontrast, high amplitude transitions, the signal E_(H) will have a highamplitude. The signal E_(H) is applied as an address to ROM 516, theoutput signal 1/D of which controls the gain, Γ, of the deemphasismultiplier 508.

The gain Γ transfer function tabulated in the LUT stored in ROM 516 ismonotonically decreasing in characteristic, as depicted by the thickline in FIG. 31 where the energy level E_(H) of the high-band luma Y_(H)applied to control signal generator 510 is shown along the horizontalaxis and the gain Γ through multiplier 508 is plotted on the verticalaxis. As may be seen in FIG. 31, as the level of the high-band lumaincreases, the gain Γ through modulator 508 correspondingly decreasesmonotonically, thus providing the deemphasis transfer function D(Y_(H))through multiplier 508. The deemphasis amount D is depicted by the thinline in FIG. 31. Optionally, for improved noise performance, at lowamplitudes of the high luma component Y_(H) a coring function may beincluded in the gain Γ transfer function tabulated in the LUT stored inROM 516 as shown by the diagonally striped area in FIG. 31. Thedeemphasis gain control signal 1/D, is generated from the memorylocation in the ROM 516 addressed by E_(H). If the measured average highluminance band energy E_(H) is very low or at zero, then the gain Γ isset near or equal to unity, passing the high luminance signal Y_(H)through multiplier 508 without any deemphasis—i. e., with noattenuation. However, when E_(H) is at a high level, the gain Γ is setat a lower value, reducing the gain through multiplier 508 below unityand thereby reducing the effective level of the luminance high-bandcomponent passed through multiplier 508 to provide maximum deemphasis.At intermediate values of E_(H) deemphasis gain control signal Γ willassume correspondingly intermediate values between zero and one, and anintermediate deemphasizing of Y_(H) will thus occur. The effect of thisadaptive deemphasis processing is to provide little or no attenuation ofthe folded luminance high-band components when they are of lowamplitude, but to provide significant attenuation when the folded lumahighs have a high amplitude. The transfer function stored as a LUT inthe ROM 516 may be such as to arrange for logarithmic compression ofhigh-band luma, for example, starting from unity gain at the coringthreshold.

FIG. 32 shows an alternative embodiment of the adaptive reemphasiscircuit 822, which is similar in its construction to the adaptivedeemphasis circuitry shown in FIG. 16. The FIG. 32 adaptive reemphasiscircuit is preferred when an adaptive deemphasis circuitry as shown inFIG. 16 is used in the record section, particularly if the adaptivedeemphasis circuitry is implemented as shown in FIG. 30. The FIG. 32adaptive reemphasis circuit is also preferred when adaptive deemphasisand folding circuitry as shown in FIGS. 34 and 35 hereinafter describedis used in the record section. The FIG. 32 adaptive reemphasis circuitmay advantageously share many common elements with the adaptivedeemphasis circuitry or the adaptive deemphasis and folding circuitry.

An input terminal 1100 of the FIG. 32 adaptive reemphasis circuitreceives the post-filtered unfolded luminance signal Y_(D)* from theoutput of soft switch 914 shown in FIGS. 9 and 10. The signal Y_(D)* isapplied to a band-splitting filter with a 2.5 MHz crossover frequency,structurally corresponding to the filters 502 and 504 of FIG. 16. FIG.32 shows the band-splitting filter configured as in FIG. 34; i. e., as ahorizontal lowpass filter (HLPF) 1102 with a 2.5 MHz break frequency anda horizontal highpass filter (HHPF) 1104 with a corresponding 2.5 MHzbreak frequency. The HHPF 1104 responds to the signal Y_(D)* to separatea recovered deemphasized high-band luminance component Y_(HD)*, whichcorresponds to the deemphasized high-band luminance component Y_(HD)output at the multiplier 508 in the FIG. 16 adaptive deemphasiscircuitry during encoding. The deemphasized high-band luminancecomponent Y_(HD)* is applied to the data input of a reemphasismultiplier 1108 structurally corresponding to the deemphasis multiplier508 in the FIG. 16 adaptive deemphasis circuitry. Y_(HD)* is alsosupplied to the input of a control signal generator 1110. Control signalgenerator 1110 includes in cascade a rectifier 1112, a horizontal lowpass filter (HLPF) 1114 and a ROM 1116 storing a look-up table (LUT),similar to the cascaded rectifier 512, HLPF 514 and ROM 516 in thecontrol signal generator 510 of FIG. 16 and FIG. 30. However, thecharacteristics of the LUT stored in ROM 1116 for use in the reemphasisprocessing differ from the characteristics of the LUT stored in ROM 516used in the deemphasis processing.

FIG. 33 shows characteristics of the LUT stored in ROM 1116 that arecomplementary to the FIG. 31 LUT characteristics that preferably arestored in ROM 516. The action of control signal generator 1110corresponds to that of control signal generator 510 in FIG. 30, exceptthat the respective transfer functions of these generators aresubstantially inverse to one another, as may seen from comparing thecharacteristic graphs in FIGS. 31 and 33.

That is, whereas the deemphasis gain Γ transfer function of the LUTstored in ROM 516 is preferably monotonically decreasing for providingluma deemphasis as shown in FIG. 31, the reemphasis gain Γ* transferfunction of the LUT stored in ROM 1116 is preferably monotonicallyincreasing as shown in FIG. 33, for providing unity gain throughreemphasis multiplier 1108 at lower signal levels of Y_(HD)* and highergain through reemphasis multiplier 1108 at higher signal levels ofY_(HD)*. By way of more specific example, when during recording thetransfer characteristic described by the LUT stored in ROM 516 providesfor logarithmic compression of the high-band luma above a threshold,during playback the transfer function stored characteristic described bythe LUT stored in ROM 1116 provides for exponential expansion of thehigh-band luma above that threshold.

Thus, reemphasis circuit 822 has the adaptive characteristic that inbroad flat image areas it provides little or no gain to the deemphasizedhigh-band luma signal Y_(HD)*. This has the resultant effect that ahigh-frequency luminance component, which was originally at a low levelin the wideband video signal input to the encoder (that is, in high-bandluma signal Y_(H) in the adaptive deemphasis and folding circuitry 108)and therefore did not undergo deemphasis during the encoding process, isnot subjected to reemphasis during the decoding processing, but isinstead passed through multiplier 1108 at substantially unity gain to besupplied at its original amplitude. On the other hand, for thoseportions of the deemphasized high-band luma signal Y_(HD)* whichcorrespond to high-frequency, high-amplitude transitions in the originalinput video signal (that is, in high-band luma signal Y_(H)) and weretherefore deemphasized during encoding, the gain through multiplier 1108is increased to restore these high-frequency luminance components totheir original level.

The adaptive reemphasis is accomplished by measuring the average energylevel (i. e., the average “local” energy) in the deemphasized high-bandluma signal Y_(HD)* by operation of rectifier 1112 and LPF 1114 toderive the average energy signal E_(H)* which is applied as an inputaddress to ROM 1116. ROM 1116 then reads out a reemphasis gain controlsignal Γ* to the gain control data input of multiplier 1108 to controlthe gain through multiplier 1108, and thereby the amount R of reemphasisperformed on the deemphasized high-band luma signal Y_(HD)*. Therelation between E_(H)*, the gain Γ* through multiplier 1108 and thereemphasis amount R is shown in FIG. 33, wherein the gain Γ* is depictedby a heavy line and the reemphasis amount R is shown by a thin line.

Referring back to FIG. 32, the reemphasized unfolded high-frequencyluminance component signal Y_(H)* output from reemphasis multiplier 1108is supplied to adder 1106 to be added together with the unfoldedlow-frequency luminance component Y_(L)* , and adder 1106 outputs thereconstructed baseband luminance signal Y* with proper amplituderelationship restored thereto and corresponding to the full-bandwidthluminance signal Y in the encoder. The reconstructed baseband luminancesignal Y* is supplied to the luminance input of composite video signalgenerator 810.

In practice, on the recording side, some amplitude boosting of very lowamplitude high-frequency luminance signal components may be done,allowing for some compression during playback, thus improving the S/Nratio in broad flat areas of the image without degrading backwardcompatibility of the encoded recorded signal. Correspondingly, in thereemphasis circuit 822 of FIG. 33, the control signal generator 1110 mayprovide a coring function, as by incorporating such a coring function inthe transfer characteristic stored in the ROM 1122 as a LUT, as depictedby the shaded area in FIG. 34.

The transfer functions stored in LUT form in the ROMs 516 and 1116 maybe modified so that the coring used to suppress noise takes place atlower levels than the coring used to set the level at which deemphasisof highband luma obtains, just as was done in the FIG. 28 reemphasiscircuit. The FIG. 31 transfer function used during recording is modifiedto insert a unity-gain “plateau” extending from the noise coring regionto the level at which deemphasis of highband luma obtains. The FIG. 33transfer function used during playback is also modified to insert aunity-gain “plateau” extending from the noise coring region to the levelat which deemphasis of highband luma obtains.

The types of deemphasis and folding circuitry 108 thusfar described aretypes in which the deemphasis of the luma high frequencies is carriedout completely before proceeding with the folding of the luminancespectrum. In a first spectrum folding technique described by Howson andBell, the video luminance signal spectrum is divided into two equalhalf-bands by band-splitter filtering, and the upper half-band is usedto modulate a sub-carrier which has its frequency set to be near theupper frequency limit of the normal video band. The reversed-spectrumlower sideband of the modulator output is selected and combined with theoriginal lower half-band to generate the folded-spectrum signal. Howsonand Bell disfavored this type of spectrum folding because of itsrequirements for complementary lowpass and bandpass filters to carry outthe initial band-splitting. However, since band-splitting is used anywayfor carrying out the deemphasis of the luma high frequencies, theinventors perceived that variants of the first spectrum-foldingtechnique described by Howson and Bell might actually prove to be atleast as practical as the second spectrum-folding technique described byHowson and Bell, espoused by them, and later adopted by Faroudja.Indeed, such variants might be more practical.

The inventors note that, for reasons of phase linearity, thecomplementary lowpass and bandpass filters used for band-splitting inthe first spectrum-folding technique described by Howson and Bell shouldbe constructed as tapped-delay-line filters if band-splitting of analogsignals is to be done. When constructed of reactive elements,complementary lowpass and bandpass analog filters respectively exhibitsubstantial lag and substantial lead at crossover frequency, making itdifficult to achieve correct mid-band phasing when the folded-spectrumluminance signal is unfolded to full bandwidth.

The inventors point out that the complementary lowpass and bandpassfilters avoided by Howson and Bell in analog design work have digitalhomologs that provide the desired complementary lowpass and bandpassresponses, but are inexpensive to construct.

In both its lowpass and bandpass responses a digital band-splittingfilter of finite-impulse-response (FIR) type exhibits linearity of phaseresponse with frequency that gives rise to uniform group delay.Accordingly, the phases of the lowpass and bandpass responses of FIRdigital filters are the same at the crossover frequency between thoseresponses.

FIG. 34 is a more detailed block diagram of an embodiment of thedeemphasis and folding circuitry 108 illustrated in FIG. 2. Thisdeemphasis and folding circuitry 108 band-splits the motion-adaptivelyspatio-temporally processed luminance signal in connection withdeemphasizing, preferably performed adaptively, of the amplitude of thehigh-frequency luminance components and folds the spectrum of thedeemphasized high-frequency luminance components into the spectrum ofthe low-frequency luminance components to produce the band-limitedluminance signal for recording. As will be more fully described below,deemphasis and folding circuitry 108 performs band splitting of theluminance signal into low-frequency components and high-frequencycomponents, appropriately de- emphasizes the high-frequency luminancecomponents, and folds the high-frequency luminance components into thespectral bandwidth of the low-frequency luminance components thereby tocompress the full-bandwidth luminance information of the input videosignal into a narrow frequency band corresponding to the bandwidth ofthe narrow band video medium.

In the deemphasis and folding circuitry 108 shown in FIG. 34, an inputterminal 501 is coupled to the output terminal of the adaptive luminanceseparator 104 of FIG. 2, that is, to the spatio-temporally processedluminance signal Y output from terminal 215 of soft switch 214 in FIG.5. Input terminal 501 receives the baseband motion-adaptivelyspatio-temporally processed luminance signal Y output from soft switch214 and couples it to an input terminal of a horizontal lowpass filter(HLPF) 502 as well as an input terminal of a horizontal highpass filter(HHPF) 504. HLPF 502 and HHPF 504 may be designed to have their −6 dBpoints each at approximately 2.5 MHz to correspond to one-half thefolding frequency. HLPF 502 and HHPF 504 are then configured to form aband-splitting filter with a 2.5 MHz crossover frequency, with HLPF 502supplying the low-frequency luminance component (low-band luma) signalY_(L) below 2.5 MHz and with HHPF 504 supplying the high-frequencyluminance component (high-band luma) signal Y_(H) above 2.5 MHz. Thelow-band luma signal Y_(L) output of HLPF 502 is coupled to a firstinput terminal of an adder 506. The high-band luma signal Y_(H) outputof HHPF 504 is coupled to a signal input terminal of a first multiplier508 which performs a deemphasis operation thereon, and also to an inputterminal of a control signal generator 510 which controls the operationof the deemphasis multiplier 508. The control signal generator 510generates a deemphasis gain control signal, which is coupled to a gaindata input terminal of deemphasis multiplier 508. The control signalgenerator 510 by way of example can be of the type shown in FIG. 30,with a ROM 516 therein storing a look-up table for the function Γ asshown in FIG. 31. The deemphasis amount D corresponds inversely to theamount of gain Γ through multiplier 508 for Y_(H). i.e., D=Γ⁽⁻¹⁾, andΓ=D⁽⁻¹⁾. Control signal generator 510 and deemphasis multiplier 508 forma deemphasis section for deemphasizing, that is, attenuating, theamplitude of the high-frequency luminance component signal Y_(H).Deemphasis processing of high-band luma Y_(H) in multiplier 508 producesa deemphasized high-frequency luminance signal Y_(HD). An amplitudemodulator 524 is used to fold the frequency spectrum of Y_(HD) over infrequency, to form a reversed frequency spectrum translated in frequencyso as to reside in a low-band extending downward from 2.5 MHz. Thisreversed frequency spectrum is suitable for spectrally interleaving withthe low-band luminance signal Y_(L) A modulation clock input terminal526 of the amplitude modulator 524 is coupled to a source (not shown) ofa folding carrier signal having a frequency 2f_(F) that is at the top ofthe frequency spectrum of Y_(HD) (e. g., about 5 MHz), which foldingcarrier signal is phase-shifted from field to field to implement a formof amplitude modulation referred to as 4-field offset modulation. Thefolding carrier signal is modulated by the deemphasized high-bandluminance output signal Y_(HD) supplied to a data input terminal of thefolding modulator 524 thereby generating a shifted-in-frequencydeemphasized high-band luminance signal Y_(HDF) with a reversedfrequency spectrum residing in a low-band extending downward from 2.5MHz. The shifted deemphasized highband luminance signal Y_(HDF) is thensupplied to the other input of adder 506, to be added back into thebaseband of the low-frequency luminance component Y_(L) to therebyproduce the interleaved band-limited folded luminance signal Y_(F) (e.g., having a bandwidth 2.5 MHz) that can be accommodated by the narrowluminance component recording bandwidth of a conventional VCR (e. g., aVHS VCR).

In digital circuitry embodying the invention in certain of its aspects,the folding carrier of frequency 2f_(F) is supplied to the amplitudemodulator 524 in digitized, sampled-data form. Modulator 524 may be astandard four quadrant multiplier, or preferably, if the samplingfrequency f_(S) is properly selected, a +1, −1 type modulator. Since thesignal Y_(HD) is randomly phased respective to the folding carrier offrequency 2f_(F), in order to meet the Nyquist sampling criterion in theamplitude modulator 524 a sampling rate f_(S) of at least 4f_(F) isrequired for the signal Y_(HD) and the folding carrier. Choosing thesampling rate f_(S) to be exactly 4f_(F) allows the folding of thedeemphasized high luminance signal Y_(HD) about a folding frequencyf_(F) one-half the folding carrier frequency 2f_(F) to be carried out inmodulator 524 by a +1, −1 type modulation operation. A +1, −1 typemodulator modulates a sampled signal by a frequency equal to one-halfthe sampling frequency by arithmetically negating every other sample.For example, if the sampling frequency is selected to be at about 10MHz, then the folding frequency will be about 5 MHz, with the actualfrequency selected so as to satisfy the above criteria relating tovertical and temporal spectral distance from vertical and temporal DC.The output signal contains a component of one-half the samplingfrequency, and upper and lower sidebands centered around +½ and −½ thesampling frequency containing the spectral information contained in theinput signal. Thus the +1, −1 amplitude modulation will shift (i. e.,alias) the high-band luma Y_(H) to a −½ lower sideband in the 2.5 MHzbandwidth of the low-band luma Y_(L).

As shown in FIG. 35, the amplitude modulator 524 of FIG. 34 may beimplemented using a multiplexer (MUX) 528 and an arithmetic negator 530.The signal Y_(HD) supplied by the data output terminal of the deemphasismultiplier 508 is applied both to an input terminal of the amplitudemodulator 524 and to a first data input terminal of the MUX 528. Anoutput terminal of the arithmetic negator 530 is coupled to a seconddata input terminal of the multiplexer 528. An output terminal of themultiplexer 528 is coupled to an input terminal of the adder 506. Afolding clock signal, which has a frequency equal to one-half thesampling clock frequency, is coupled to the control input terminal 532of the multiplexer. This signal alternates between a logic “1” value anda logic “0” value at the sampling frequency, and may be generated by aflip-flop coupled to the sampling clock signal.

In operation, when the folding clock signal is a logic “1” signal, thenthe multiplexer 528 couples the non-negated (+1) signal from the inputterminal to its output terminal. When the folding clock signal is alogic “0” signal, then the multiplexer couples the negated (+1) signalfrom the arithmetic negator 530 to its output terminal. In this fashion,a (+1, −1) modulated signal is reproduced. The lower sideband of themodulated signal contains a spectral image of the deemphasized high-bandluminance signal Y_(HD) that is in reversed spectrum form. That is, thedeemphasized high-band luminance signal Y_(HD) is folded about thefolding frequency such that the lower frequency components of thedeemphasized luminance high-band frequencies are folded into thebandwidth below 2.5 MHz, and the higher frequency components of thedeemphasized high-band luminance frequencies of 4.2 MHz, for example,are folded into the neighborhood of 800 kHz, thus producing the foldeddeemphasized high-band luminance signal Y_(HDF).

The folded deemphasized high-band luminance signal Y_(HDF) is thencombined with the low-band luminance signal Y_(L) in the adder 506. Theadder 506 supplies the composite folded luminance signal Y_(F) whichcontains the luminance information of the input luminance wide basebandsignal Y compressed within a folded bandwidth of 2.5 MHz, thus making itpossible to transmit the 4.2 MHz NTSC baseband luminance information viaa narrow 2.5 MHz bandwidth medium such as by a conventional narrowbandwidth format VCR and videocassette.

The folded luminance signal Y_(F) may then be supplied to a recordequalization section 522, as shown in FIG. 35, where the signal isequalized prior to digital-to-analog conversion, to pre-compensate forloss in the tape path and to compensate for encoder processing losses.This equalization boosts the frequencies around the 2.5 MHz region tocompensate for the signal attenuation characteristic in the band splitregion of the deemphasis circuit band splitting filter, for example. Thefolded luminance signal Y_(F) from the folding section 108 is thensupplied to the DAC 110 as shown in FIG. 2 to be converted to an analogluminance signal Y_(R) that frequency modulates the luma carrier, and isultimately recorded onto videotape by the VCR 40 of FIG. 1.

FIG. 36 shows a modification of the FIG. 34 deemphasis and foldingcircuitry, in which modification elements corresponding to those in FIG.34 are designated with like reference numerals. In FIG. 34 the luminancehigh-band signal Y_(H) is adaptively deemphasized prior to foldingmodulation and addition with the lowband luma signal Y_(L). In the FIG.36 deemphasis and folding circuitry an equivalent result is effected byreversing the order of the multiplier 508 and modulator 524 from thatshown in FIG. 34. In FIG. 36 the luma high-band signal Y_(H) is foldedin the folding modulator 524 and then the folded high-band output signaY_(HF) from the folding modulator 524 is adaptively deemphasized in thedeemphasis multiplier 508.

In the combined adaptive deemphasis and folding circuitry shown in FIGS.34 and 36 the modulator 524 performs the folding operation only upon thehigh-band luminance component signal. The image of the high-bandluminance component signal folded into the DC to 2.5 MHz frequency rangeas a reversed spectrum has an image in the 7.5-10 MHz range, which imagehas a non-reversed spectrum.

The 3-8 MHz range is essentially free of signal. Referring back to FIG.2, the image in the 7.5-10 MHz range is suppressed by the sampling clockrejection lowpass filter of the digital-to-analog converter 110preceding the VCO 120 that modulates the frequency of the luma carrierwith the folded luminance signal, assuming the digital sampling rate ischosen to be about 10 MHz. Unless one wishes to suppress noise in therange immediately above 2.5 MHz in the output signal from the foldingsection 108, it is not necessary to follow the adder 506 with a lowpassfilter cutting off at the folding frequency. The cut-off frequency ofthe sampling clock rejection lowpass filter associated with the DAC 110can be below the 2f_(F) folding carrier frequency so as to suppress thatcarrier (or remnants of it where the carrier is suppressed by themodulator 524 being of balanced nature), but at the same time being wellabove the 2.5 MHz folding frequency f_(F) so as not to lose energy inthe region of the folding frequency f_(F). This contrasts with thesecond type of folding procedure as described by Howson and Bell and asimplemented both by them and by Faroudja where a lowpass filter cuttingoff at the folding frequency f_(F) is used. As will be described below,the second type of folding procedure can be modified so the lowpassfilter cuts off at a frequency somewhat higher than the foldingfrequency f_(F), but it is desirable that a lowpass filter be used afterthe combining of the normal-spectrum and reversed-spectrum luma signalsor artifacts caused by spurious horizontal high frequencies arenoticeable in an image played back from tape.

In the folding circuit embodiments shown in FIGS. 34 and 36 employing inthe deemphasis processing a band-splitting filter as thusfar described,the bandwidth of the compatibly-recorded luminance signal Y_(R) extendsonly to around 2.5 MHz (i. e., the upper limit of the low-band luma outof the band splitting filter), while the luma frequencies above 2.5 MHzare carried in the folded signal. This limiting of the recordedluminance bandwidth is of no significant consequence when the recordedsignal is reproduced by a playback apparatus including (in accordancewith an aspect of the present invention) a decoder for unfolding thecompatibly encoded recorded folded luminance signal and reconstructing awide bandwidth luminance signal therefrom. In such playback apparatusthe folded luminance frequencies extending beyond 2.5 MHz are recoveredin playback decoding, so as to display an image having full horizontalresolution. However, when playing back the compatibly encoded recordingon a conventional playback apparatus lacking such a decoding facility,the displayed horizontal resolution is limited by the limited bandwidthof the reproduced luminance signal, since the higher luma frequenciescarried in the folded signal are not recovered.

FIG. 37 shows a modification that can be made to the FIG. 34 or FIG. 36adaptive deemphasis and folding circuit. The HLPF 502 and HHPF 504forming the band-splitting filter in FIGS. 34 and 36 are replacedrespectively by a horizontal lowpass filter (HLPF) 1502 which may haveits characteristics selected to provide a −6 dB response at around 3MHz, and a vertical highpass filter (VHPF) 1504 providing a −6 dBresponse at 2 MHz, with both HLPF 1502 and VHPF 1504 receiving the inputluminance signal Y. HLPF 1502 and HHPF 1504 together perform aband-splitting function, however, their respective output signals Y_(L)′and Y_(H)′ are not substantially within abutting half- or split-bandsbut rather are in respective bands that substantially overlap each otherin frequency by 1 MHz or so. The high-band luma signal Y_(H)′ from HHPF1504 is adaptively deemphasized and heterodyned with the folding carrierby adaptive deemphasis and folding circuitry 1560, which can take a formsimilar to that in FIG. 34 or in FIG. 36.

Circuitry 1560 responds to the high-band luma signal Y_(H)′ to provide aband-shifted deemphasized high-frequency luma signal Y_(HFD)′ to theadder 1506 for combining with the low-band luma signal Y_(L)′. The sumoutput signal from the adder 1506 is a folded luma signal Y_(F)′supplied to VCR 40 of FIG. 1.

Because the low-band luminance signal Y_(L)′ from HLPF 1502 has abandwidth extending up to approximately 3 MHz, so also the foldedluminance signal Y_(F)′ will have a bandwidth extending to approximately3 MHz at the high end. i.e., its frequency characteristic will be 6 dBdown at 3 MHz, thereby providing the advantage of the folded luminancesignal Y_(F)′ having approximately 0.5 MHz greater bandwidth over thefolded luminance signal Y_(F) output by the folding circuitry previouslydescribed. Thus, the recorded luminance signal Y_(R)′ will contain thelow-band luminance components up to 3 MHz, as well as the band shiftedhigh luminance frequencies above 2 MHz folded within the limitedbandwidth occupied by the low-frequency luminance components.Accordingly, the folded limited-bandwidth luminance signal Y_(F), whenrecorded and then reproduced by conventional narrow bandwidth playbackapparatus lacking a facility for recovering the folded high-frequencyluminance component will provide greater horizontal resolution than willthe folded limited-bandwidth luminance signal Y_(F) provided by thefolding circuitry previously described, offering the advantage of higherhorizontal resolution in compatible playback.

Additional advantages of the use of HLPF 1502 and VHPF 1504 in thefolding circuit of FIG. 37 are that both the low and high bands havefull energy in the 2.5 MHz region, reducing or avoiding the need forequalization of the folded luminance signal Y_(F)′ prior to recording.Having both the low and high bands have full energy in the 2.5 MHzregion is desirable when unfolding during playback. The folded luminancesignal Y_(F)′ can be lowpass filtered with an FIR filter 6 dB down at2.5 MHz, then unfolded without incurring in the 2.5 MHz crossover regiona dip in energy or a distortion of phasing.

While HLPF 1502 and VHPF 1504 can be constructed so as to share the sametapped delay line for the folded luminance signal Y_(F)′ the two filterswill have to use different-amplitude weighting coefficients; so theirweighted summation networks will, at least for the most part, have to beseparate from each other.

FIG. 38 shows adaptive deemphasis processing according to an aspect ofthe present invention being applied to a folding system in which foldingis performed on the baseband luminance signal Y, as in the second typeof folding procedure described by Howson and Bell. The basebandluminance signal Y is applied to one input of an adder 550 and also toan adaptive deemphasis circuit 560, where the high-frequency luminancecomponent is adaptively deemphasized with a monotonically decreasingtransfer function. The deemphasized baseband luminance signal is thenapplied to folding modulator 570, where it is shifted as by +1, −1multiplexing in accordance with a folding clock as described above, andthe deemphasized shifted baseband luminance signal is then applied tothe other input of adder 550 to be combined with the input basebandluminance signal. The interleaved luminance signal from adder 550 isthen passed through a horizontal lowpass filter 580, then converted toanalog form and recorded as previously described. The HLPF 580 can bedesigned to have a cut-off frequency just above the folding frequencyf_(F) of 2.5 MHz as described by Howson and Bell or by Faroudja.

However, it is preferred by the present inventors to extend the responseof the HLPF 580 so the recorded luminance signal Y_(R)′ will contain thelow-band luminance components up to 3 MHz, as well as the band shiftedhigh luminance frequencies above 2 MHz folded within the limitedbandwidth occupied by the low-frequency luminance components. Thisoffers the advantage of higher horizontal resolution in compatibleplayback on a conventional VHS VCR or VCP. Also, when playing back on aVCR or VCP having unfolding circuitry, the folded luminance signalY_(F)′ can be lowpass filtered with an FIR filter 6 dB down at 2.5 MHz,then unfolded without incurring in the 2.5 MHz crossover region a dip inenergy or a distortion of phasing.

A number of other folding and unfolding circuits suitable for usetogether with the adaptive deemphasis and reemphasis circuitry embodyingaspects of the invention are described by C. H. Strolle et alii in U.S.patent application Ser. No. 819,890 filed Jan. 13, 1992. entitledDIGITAL MODULATORS FOR USE WITH SUB-NYQUIST SAMPLING OF RASTER-SCANNEDSAMPLES OF IMAGE INTENSITY, and incorporated herewithin.

Alternatively, folded-spectrum signals can be formed by subtractivelycombining a reversed-frequency-spectrum signal with anon-reversed-frequency-spectrum, or normal-frequency-spectrum, signal aswell as by additively combining such signals. Reversing of the polarityof the signal modulating the folding carrier may alternatively oradditionally be done. Reversing of the polarity of the signal thefolding carrier may alternatively or additionally be done. These variousprocedures only invert the phase of the reversed-frequency-spectrumsignal and do not affect the amplitude of its frequency spectrum. Asimilar variety of ways to invert the phase of thereversed-frequency-spectrum signal during unfolding also are possible.Modifications of this sort of the embodiments of the invention describedabove are considered to be further embodiments of the invention. In theclaims which follow, the term “means for linearly combining” is to beregarded to be a generic term including within its scope both adders andsubtractors, with or without weighting of signal inputs before adding orsubtracting.

Deemphasis and subsequent reemphasis of high-energy reversed-spectrumcomponents of the reversed-spectrum components of a folded video signalis described above with particular attention to a modified videorecording/playback system for recording magnetic tape cassettes insubstantial accordance with the VHS standard. One skilled in the art andacquainted with the foregoing disclosure will be enabled to design othervideo processing systems improved by embodying the invention in itsvarious aspects, and this should be borne in mind when construing thescope of the claims which follow.

Appendix

The specification and drawing of U.S. patent application serial No.008,813 filed Jan. 25, 1993 by C. H. Strolle et alii, entitled ADAPTIVEDEEMPHASIS AND REEMPHASIS OF HIGH FREQUENCIES IN VIDEO TAPE RECORDING,UTILIZING A RECORDED CONTROL SIGNAL, and assigned to Samsung ElectronicsCo., Ltd., are appended hereto for purposes of incorporation into theforegoing specification.

The specification and drawing of U.S. patent application serial No.819,890 filed Jan. 13, 1992 by C. H. Strolle et alii, entitled DIGITALMODULATORS FOR USE WITH SUB-NYQUIST SAMPLING OF RASTER-SCANNED SAMPLESOF IMAGE INTENSITY, and assigned to Samsung Electronics Co., Ltd., areappended hereto for purposes of incorporation into the foregoingspecification.

The specification and drawings of U.S. patent application Ser. No.08/059,765 filed May 11, 1993 by C. H. Strolle, et alii entitledFREQUENCY-MULTIPLEXING FM LUMA SIGNAL WITH COLOR AND 2ND UNDER SIGNALSHAVING OVERLAPPING FREQUENCY SPECTRA, and assigned to SamsungElectronics Co., Ltd., are appended hereto for purposes of incorporationinto this specification.

The specification and drawings of U.S. patent application serial No.07/787,690 filed Nov. 4, 1991 by C. H. Strolle, et alii entitled SYSTEMFOR RECORDING AND REPRODUCING A WIDE BANDWIDTH VIDEO SIGNAL VIA A NARROWBANDWIDTH MEDIUM, and assigned to Samsung Electronics Co., Ltd., areappended hereto for purposes of incorporation into this specification.

The specification and drawings of U.S. patent application Ser. No.07/604,493 filed Oct. 26, 1990 by C. H. Strolle, et alii entitledADAPTIVE DEEMPHASIS AND REEMPHASIS OF HIGH FREQUENCIES IN A VIDEOSIGNAL, and assigned to Samsung Electronics Co., Ltd., are appendedhereto for purposes of incorporation into this specification.

What is claimed is:
 1. Apparatus for reversing the spectrum of a videosignal descriptive of the scanning of successive image frames suppliedat a prescribed frame rate, which scanning is done line-by-line at aprescribed scan line rate, said apparatus comprising: a source of saidvideo signal having a lower video frequency portion and an upper videofrequency portion; a generator of a carrier signal of a carrierfrequency that is a multiple of said prescribed scan line rate and abovesaid upper video frequency portion; a multiplier having a multiplicandinput terminal coupled to receive one of said video and carrier signalsand a multiplier input terminal coupled to receive another of said videoand carrier signals for generating a product signal having a firstamplitude-modulation sideband that is lower in frequency than saidcarrier frequency and having a second amplitude-modulation sideband thatis higher in frequency than said carrier frequency; and means forreversing at said prescribed scan line rate the phase of the carriersignal received by said multiplier, each reversal being at a respectiveinstant between successive scan lines, thereby enabling conversion ofsaid product signal into a reversed-spectrum video signal withline-to-line inversion of phase.
 2. Apparatus as set forth in claim 1,in combination with: means for linearly combining said video signal withsaid reversed-spectrum video signal with line-to-line alternation ofphase to generate a combined signal.
 3. A combination as set forth inclaim 2, wherein said means for linearly combining said video signalwith said reversed-spectrum video signal with line-to-line alternationof phase essentially consists of: an adder connected for receiving saidvideo signal as a first summand input signal, for receiving saidreversed-spectrum video signal with line-to-line alternation as a secondsummand input signal, and for supplying said combined signal as a sumoutput signal.
 4. Apparatus as set forth in claim 1, further comprising:means for lowpass filtering said combined signal, having a bandwidththat extends only to a frequency that is not higher than said carrierfrequency.
 5. Apparatus as set forth in claim 1, further comprising:means for lowpass filtering said combined signal, having a bandwidththat extends only to a frequency that is not substantially higher thanone half said carrier frequency.
 6. Apparatus for folding the spectrumof a video signal descriptive of the scanning of successive image framessupplied at a prescribed frame rate, which scanning is done line-by-lineat a prescribed scan line rate, said apparatus comprising: a source ofsaid video signal having a bandwidth extending across baseband to anupper video frequency; a generator of a carrier signal of a carrierfrequency that is a multiple of said prescribed scan line rate and abovesaid upper video frequency; means for reversing the phase of the carriersignal at said prescribed scan line rate, each reversal being at arespective instant between successive scan lines, to supply a foldingcarrier signal; a highpass filter responsive to said video signal forsupplying a high band video signal; means for detecting the level ofenergy in said high band video signal, thereby to generate a detectedenergy signal; means responsive to said detected energy signal exceedinga prescribed threshold level, for generating a gain control signal thatis of maximum value until said prescribed threshold level is exceeded bysaid detected energy signal, that monotonically decreases in value assaid prescribed threshold level is exceeded in greater amount by saiddetected energy signal, and that is zero for said detected energy signalreaching its own maximum value; a first and a second multiplier eachhaving a respective multiplicand input terminal, a respective multiplierinput terminal and a respective product output terminal, the productoutput terminal of said first multiplier connecting to the multiplicandinput terminal of said second multiplier, said high band video signaland said gain control signal and said folding carrier signal beingapplied to separate ones of the multiplicand input terminal of saidfirst multiplier and the multiplier input terminal of said firstmultiplier and the multiplier input terminal of said second multiplier;and means for linearly combining said video signal with a product signalsupplied from the product output terminal of said second multiplier, togenerate a combined signal.
 7. Apparatus as set forth in claim 6,wherein said means for linearly combining said video signal with aproduct signal essentially consists of: an adder connected for receivingsaid video signal as a first summand input signal, for receiving saidproduct signal as a second summand input signal, and for supplying saidcombined signal as a sum output signal.
 8. Apparatus as set forth inclaim 6, further comprising: means for lowpass filtering said combinedsignal, having a bandwidth that extends only to a frequency that is nothigher than said carrier frequency.
 9. Apparatus as set forth in claim6, further comprising: means for lowpass filtering said combined signal,having a bandwidth that extends only to a frequency that is notsubstantially higher than one half said carrier frequency.
 10. Apparatusas set forth in claim 6, wherein said high band video signal is appliedto the multiplicand input terminal of said first multiplier, said gaincontrol signal is applied to the multiplier input terminal of said firstmultiplier, and said folding carrier signal is applied lo the multiplierinput terminal of said second multiplier.
 11. Apparatus as set forth inclaim 6, wherein said high band video signal is applied to themultiplicand input terminal of said first multiplier, said foldingcarrier signal is applied to the multiplier input terminal of said firstmultiplier, and said gain control signal is applied to the multiplierinput terminal of said second multiplier.
 12. Apparatus as set forth inclaim 6, wherein said means responsive to said detected energy signalexceeding a prescribed threshold level for generating a gain controlsignal comprises: a rectifier for generating a rectified response tosaid detected energy signal exceeding said prescribed threshold level; alowpass filter for lowpass filtering said rectified response to saiddetected energy signal exceeding said prescribed threshold level toprovide an average energy signal representing an average energy of saiddetected energy signal; and a look-up table for providing said gaincontrol signal in accordance with said average energy signal that is ofmaximum value until said prescribed threshold level is exceeded by saiddetected energy signal, that monotonically decreases in value as saidprescribed threshold level is exceeded in greater amount by saiddetected energy signal, and that is zero when said detected energysignal reaches a maximum value.
 13. Apparatus for folding the spectrumof a video signal descriptive of the scanning of successive image framessupplied at a prescribed frame rate, which scanning is done line-by-lineat a prescribed scan line rate, said apparatus comprising: a source ofsaid video signal having a bandwidth extending across baseband to anupper video frequency; a generator of a carrier signal of a carrierfrequency that is a multiple of said prescribed scan line rate and thatis substantially the same in frequency as said upper video frequency;means for reversing the phase of the carrier signal at said prescribedscan line rate, each reversal being at a respective instant betweensuccessive scan lines, to supply a folding carrier signal; a lowpassfilter responsive to said video signal, for supplying a low band videosignal extending up to a frequency at least as high as one half saidcarrier frequency; a highpass filter responsive to said video signal,for supplying a high band video signal extending up from a frequency atleast as low as one half said carrier frequency; means for detecting thelevel of energy in said high band video signal, thereby to generate adetected energy signal; means responsive to said detected energy signalexceeding a prescribed threshold levels for generating a gain controlsignal that is of maximum value until said prescribed threshold level isexceeded by said detected energy signal, that monotonically decreases invalue as said prescribed threshold level is exceeded in greater amountby said detected energy signal, and that is zero for said detectedenergy signal reaching its own maximum value; a first and a secondmultiplier each having a respective multiplicand input terminal, arespective multiplier input terminal and a respective product outputterminal, the product output terminal of said first multiplierconnecting to the multiplicand input terminal of said second multiplier,said high band video signal and said gain control signal and saidfolding carrier signal being applied to separate ones of themultiplicand input terminal of said first multiplier and the multiplierinput terminal of said first multiplier and the multiplier inputterminal of said second multiplier; and means for linearly combiningsaid low band video signal with a product signal supplied from theproduct output terminal of said second multiplier to generate a combinedsignal.
 14. Apparatus as set forth in claim 13, wherein said means forlinearly combining said video signal with a product signal essentiallyconsists of: an adder connected for receiving said video signal as afirst summand input signal, for receiving said product signal as asecond summand input signal, and for supplying said combined signal as asum output signal.
 15. Apparatus as set forth in claim 13 furthercomprising: means for lowpass filtering said combined signal, having abandwidth that extends only to a frequency that is not higher than saidcarrier frequency.
 16. Apparatus as set forth in claim 13, furthercomprising: means for lowpass filtering said combined signal, having abandwidth extending across baseband past one half said upper videofrequency, but not extending to said upper video frequency. 17.Apparatus as set forth in claim 13, wherein said high band video signalis applied to the multiplicand input terminal of said first multiplier,said gain control signal is applied to the multiplier input terminal ofsaid first multiplier, and said folding carrier signal is applied to themultiplier input terminal of said second multiplier.
 18. Apparatus asset forth in claim 13, wherein said high band video signal is applied tothe multiplicand input terminal of said first multiplier, said foldingcarrier signal is applied to the multiplier input terminal of said firstmultiplier, and said gain control signal is applied to the multiplierinput terminal of said second multiplier.
 19. Apparatus as set forth inclaim 13, wherein said means responsive to said detected energy signalexceeding a prescribed threshold level for generating a gain controlsignal comprises: a rectifier for generating a rectified response tosaid detected energy signal exceeding said prescribed threshold level; alowpass filter for lowpass filtering said rectified response to saiddetected energy signal exceeding said prescribed threshold level toprovide an average energy signal representing an average energy of saiddetected energy signal; and a look-up table for providing said gaincontrol signal in accordance with said average energy signal that is ofmaximum value until said prescribed threshold level is exceeded by saiddetected energy signal, that monotonically decreases in value as saidprescribed threshold level is exceeded in greater amount by saiddetected energy signal, and that is zero when said detected energysignal reaches a maximum value.
 20. Apparatus for folding the spectrumof a video signal descriptive of the scanning of successive image framessupplied at a prescribed frame rate, which scanning is done line-by-lineat a prescribed scan lire rate, said apparatus comprising: a source ofsaid video signal having a bandwidth extending across baseband to anupper video frequency; a generator of a carrier of a carrier frequencythat is a multiple of said prescribed scan line rate and that is atleast as high in frequency as said upper video frequency; means forreversing the phase of the carrier frequency at said prescribed scanline rate, each reversal being at a respective instant betweensuccessive scan lines, to supply a folding carrier; means for generatingamplitude modulation of said folding carrier in accordance with saidvideo signal; and means for linearly combining said video signal andsaid amplitude modulation of said folding carrier to generate a combinedsignal.
 21. Apparatus as set forth in claim 20, wherein said means forlinearly combining said video signal and said amplitude modulation ofsaid folding carrier consists of: an adder connected for receiving saidvideo signal as a first summand input signal, for receiving saidamplitude modulation of said folding carrier as a second summand inputsignal, and for supplying said combined signal as a sum output signal.22. Apparatus as set forth in claim 20, further comprising: means forlowpass filtering said combined signal, having a bandwidth that extendsonly to a frequency that is not higher than said carrier frequency. 23.Apparatus as set forth in claim 20, further comprising: means forlowpass filtering said combined signal, having a bandwidth extendingacross baseband past one half said upper video frequency, but notextending to said upper video frequency.
 24. Apparatus for folding thespectrum of a video signal descriptive of the scanning of successiveimage frames supplied at a prescribed frame rate, which scanning is doneline-by-line, at a prescribed scan lire rate, said apparatus comprising:a source of said video signal having a bandwidth extending acrossbaseband to an upper video frequency; a generator of a carrier of acarrier frequency that is a multiple of said prescribed scan line rateand that is at least as high in frequency as said upper video frequency;means for reversing the phase of the carrier at said prescribed scanline rate, each reversal being at a respective instant betweensuccessive scan lines, to supply a folding carrier; de-emphasis meansfor de-emphasizing said upper video frequency of said video signal andproviding said video signal having deemphasized high frequencies; meansfor generating amplitude modulation of said folding carrier inaccordance with said video signal having said deemphasized highfrequencies; and means for linearly combining said video signal and saidamplitude modulation of said folding carrier to generate a combinedsignal having a first amplitude-modulation sideband that is lower infrequency than said carrier frequency and having a secondamplitude-modulation sideband that is higher in frequency than saidcarrier frequency.
 25. Apparatus as set forth in claim 24, wherein saidde-emphasis means comprises: a band-separation filter responding to saidvideo signal for separating a lower video frequency portion and an uppervideo frequency portion from said video signal, said lower videofrequency portion comprising only spectral components in alower-frequency band extending up to said carrier frequency, and saidupper video frequency portion comprising only spectral components in anupper-frequency band extending above said carrier frequency; a rectifierfor generating a rectified response to said upper video frequencyportion separated from said video signal; a threshold circuit forresponding only to portions of said rectified response exceeding apredetermined threshold level; a lowpass filter for producing inresponse to the response of said threshold circuit a deemphasis gaincontrol signal dependent on the amount the level of said upper videofrequency portion exceeds said predetermined threshold level; meansresponsive to said deemphasis gain control signal, for generating amultiplier signal by multiplying said upper video frequency portion ofsaid video signal by said deemphasis gain control signal; and means foradditively combining said multiplied signal with said lower videofrequency portion of said video signal to provide said video signalhaving said de-emphasized high frequencies to said means for generatingamplitude modulation of said folding carrier enabling generation of saidcombined signal.
 26. Apparatus as set forth in claim 25, wherein saidband-separation filter comprises: a highpass filter for highpassfiltering said video signal to separate said upper video frequencyportion thereof; and a lowpass filter for lowpass filtering said videosignal to separate said lower video frequency portion thereof. 27.Apparatus as set forth in claim 25, wherein said means responsive tosaid deemphasis gain control signal for generating said multipliersignal essentially consists of means for subtracting said deemphasisgain control signal from a prescribed level.
 28. Apparatus as set forthin claim 25, wherein said band-separation filter comprises: alinear-phase finite-impulse-response filter comprising a tapped delayline with an odd number of taps spaced at one sample intervals, and aweighted summation circuit for weight samples with alternate negativeand positive weights to provide a symmetrical kernel indicative of ahighpass filter response of said video signal; and a subtractor forsubtracting from said video signal said upper video frequency portionseparated from said video signal to provide said lower video frequencyportion.
 29. Apparatus as set forth in claim 25, wherein saidde-emphasis means comprises: means for separating an lower videofrequency portion and an upper video frequency portion from said videosignal; means for rectifying said upper video frequency portionseparated from said video signal to generate a rectified upper videofrequency response; means for baseline clipping said rectified uppervideo frequency response to generate a deemphasis gain control signal;means for generating a multiplier signal having an amplitude that variesin response to said deemphasis gain control signal, growing smaller withincrease in said deemphasis gain control signal and growing larger withdecrease in said deemphasis gain control signal; means for additivelycombining said multiplier signal with said lower video frequency portionseparated from said video signal to provide said video signal havingsaid de-emphasized high frequencies to said means for generatingamplitude modulation of said folding carrier enabling generation of saidcombined signal.
 30. Apparatus as set forth in claim 29, wherein saidmeans for separating said lower video frequency portion and said uppervideo frequency portion from said video signal comprises: a highpassfilter for highpass filtering said video signal to generate said uppervideo frequency portion; and a lowpass filter for lowpass filtering saidvideo signal to generate said lower video frequency portion. 31.Apparatus for folding the spectrum of a video signal descriptive of thescanning of successive image frames supplied at a prescribed frame rate,which scanning is done line-by-line at a prescribed scan lire rate, saidapparatus comprising: a source of said video signal having a bandwidthextending across baseband to an upper video frequency; a generator of acarrier of a carrier frequency that is a multiple of said prescribedscan line rate and that is at least as high in frequency as said uppervideo frequency; means for reversing the phase of the carrier at saidprescribed scan line rate, each reversal being at a respective instantbetween successive scan lines, to supply a folding carrier; means forgenerating amplitude modulation of said folding carrier in accordancewith said video signal, said amplitude modulation means comprising meansfor deemphasizing said upper video frequency of said video signal togenerate a deemphasized high band video signal, and a four-quadrantmultiplier for multiplying said folding carrier with said deemphasizedhigh band video signal to generate said amplitude modulation of saidfolding carrier; and means for linearly combining said video signal andsaid amplitude modulation of said folding carrier to generate a combinedsignal.
 32. Apparatus for folding the spectrum of a video signaldescriptive of the scanning of successive image frames supplied at aprescribed frame rate, which scanning is done line-by-line at aprescribed scan lire rate. said apparatus comprising: a source of saidvideo signal having a bandwidth extending across baseband to an uppervideo frequency; a generator of a carrier of a carrier frequency that isa multiple of said prescribed scan line rate and that is at least ashigh in frequency as said upper video frequency; means for reversing thephase of the carrier at said prescribed scan line rate, each reversalbeing at a respective instant between successive scan lines, to supply afolding carrier; means for generating amplitude modulation of saidfolding carrier in accordance with said video signal, said amplitudemodulation means comprising: means for detecting the energy level ofupper video frequency of said video signal to generate a detected energysignal; means responsive to said detected energy signal exceeding aprescribed threshold level, for generating a gain control signal that isof maximum value until said prescribed threshold level is exceeded bysaid detected energy signal, that monotonically decreases in value assaid prescribed threshold level is exceeded in greater amount by saiddetected energy signal, and that is zero when said detected energysignal reaches its own maximum value; a first four-quadrant multiplierfor multiplying said gain control signal with said upper video frequencyof said video signal to generate a deemphasized high band video signal;and a second four-quadrant multiplier for multiplying said foldingcarrier with said deemphasized high band video signal to generate saidamplitude modulation of said folding carrier; and means for linearlycombining said video signal and said amplitude modulation of saidfolding carrier to generate a combined signal.
 33. Apparatus for foldingthe spectrum of a video signal descriptive of the scanning of successiveimage frames supplied at a prescribed frame rate, which scanning is doneline-by-line at a prescribed scan line rate, said apparatus comprising:a source of said video signal having a bandwidth extending acrossbaseband to an upper video frequency; a generator of a carrier frequencythat is a multiple of said prescribed scan line rate and that is atleast as high in frequency as said upper video frequency; means forreversing the phase of the carrier frequency at said prescribed scanline rate, each reversal being at a respective instant betweensuccessive scan lines, to supply a folding carrier; a highpass filterresponsive to said video signal, for supplying a high band video signal;means for detecting the level of energy in said high band video signal,thereby to generate a detected energy signal; means responsive to saiddetected energy signal exceeding a prescribed threshold level forgenerating a gain control signal that is of maximum value until saidprescribed threshold level is exceeded by said detected energy signal,that monotonically decreases in value as said prescribed threshold levelis exceeded in greater amount by said detected energy signal, and thatis zero for said detected energy signal reaching its own maximum value;means for generating amplitude modulation of said folding carrier inaccordance with said video signal; means for regulating said amplitudemodulation of said folding carrier in accordance with said gain controlsignal, included in said means for generating amplitude modulation ofsaid folding carrier in accordance with said video signal; and means forlinearly combining said video signal and said amplitude modulation ofsaid folding carrier to generate a combined signal.
 34. Apparatus forfolding the spectrum of a video signal descriptive of the scanning ofsuccessive image frames supplied at a prescribed frame rate, whichscanning is done line-by-line at a prescribed scan line rate, saidapparatus comprising: a source of said video signal having a bandwidthextending across baseband to an upper video frequency; a generator of acarrier frequency that is a multiple of said prescribed scan line rateand that is at least as high in frequency as said upper video frequency;means for reversing the phase of the carrier frequency at saidprescribed scan line rate, each reversal being at a respective instantbetween successive scan lines, to supply a folding carrier; a lowpassfilter responsive to said video signal, for supplying a low band videosignal extending up to a frequency at least as high as one half saidcarrier frequency; a highpass filter responsive to said video signal,for supplying a high band video signal extending up from a frequency atleast as low as one half said carrier frequency; means for generatingamplitude modulation of said folding carrier in accordance with saidhigh band video signal; and means for combining said low band videosignal with at least the portion of said amplitude modulation of saidfolding carrier extending up to a frequency at least as high as one halfsaid carrier frequency, thereby providing a folded-spectrum videosignal.
 35. Apparatus as set forth in claim 34, wherein said means forgenerating amplitude modulation of said folding carrier in accordancewith said high band video signal comprises: means for deemphasizing saidhigh band video signal to generate a deemphasized high band videosignal; and a four-quadrant multiplier for multiplying said foldingcarrier with said deemphasized high band video signal to generate saidamplitude modulation of said folding carrier.
 36. Apparatus as set forthin claim 34, wherein said means for generating amplitude modulation ofsaid folding carrier in accordance with said high band video signalcomprises: means for detecting the level of energy in said high bandvideo signal to generate a detected energy signal; means responsive tosaid detected energy signal exceeding a prescribed threshold level forgenerating a gain control signal that is of maximum value until saidprescribed threshold level is exceeded by said detected energy signal,that monotonically decreases in value as said prescribed threshold levelis exceeded in greater amount by said detected energy signal, and thatis zero for said detected energy signal reaching its own maximum value;a first four-quadrant multiplier for multiplying said gain controlsignal with said high band video signal to generate a deemphasized highband video signal; and a second four-quadrant multiplier for multiplyingsaid folding carrier with said deemphasized high band video signal togenerate said amplitude modulation of said folding carrier. 37.Apparatus for folding the spectrum of a full-band video signaldescriptive of the scanning of successive image frames supplied at aprescribed frame rate, which scanning is done line-by-line at aprescribed scan line rate, said apparatus comprising: a source of saidfull-band video signal having a bandwidth extending across baseband toan upper video frequency; a generator of a carrier frequency that is amultiple of said prescribed scan line rate and that is at least as highin frequency as said upper video frequency; means for reversing thephase of the carrier frequency at said prescribed scan line rate, eachreversal being at a respective instant between successive scan lines, tosupply a folding carrier; a band-splitting filter responsive to saidfull-band video signals for supplying a low band video signal and a highband video signal respectively substantially below and substantiallyabove a crossover frequency that is one half said carrier frequency;means for adaptively deemphasizing said high band video signal togenerate a deemphasized high band video signal; means for combining saiddeemphasized high band video signal and said low band video signal togenerate a modified full-band video signal; means for generatingamplitude modulation of said folding carrier in accordance with saidmodified full-band video signal; means for combining said modifiedfull-band video signal with said amplitude modulation of said foldingcarrier to generate a combined signal; and a lowpass filter forreceiving said combined signal and for supplying response to the portionof said combined signal that is below said crossover frequency, andsuppressing the portion of said combined signal that is above saidcrossover frequency.
 38. Apparatus as set forth in claim 37, whereinsaid means for adaptively deemphasizing said high band video signal togenerate said deemphasized high band video signal comprises: a rectifierfor generating a rectified response to said high band video signal; athreshold circuit for responding only to portions of said rectifiedresponse exceeding a predetermined threshold level; a lowpass filter forgenerating said deemphasis gain control signal by lowpass filtering saidhigh band video signal, said deemphasis gain control signal being variedin response to the energy level of said high band video signal exceedingsaid predetermined threshold level; means responsive to said deemphasisgain control signal, for generating a multiplier signal by multiplyingsaid high band video signal by said deemphasis gain control signal; andmeans for additively combining said multiplied signal with said low bandvideo signal to provide said modified full-band video signal. 39.Apparatus as set forth in claim 38, wherein said band-splitting filtercomprises: a highpass filter for highpass filtering said full-band videosignal to generate said high band video signal above said crossoverfrequency that is one half said carrier frequency; and a lowpass filterfor lowpass filtering said full-band video signal to generate said lowband video signal below said crossover frequency that is one half saidcarrier frequency.
 40. Apparatus for folding the spectrum of a videosignal descriptive of the scanning of successive image frames suppliedat a prescribed frame rate, which scanning is done line-by-line at aprescribed scan line rate, said apparatus comprising: a source of saidvideo signal having a bandwidth extending across baseband to an uppervideo frequency; a generator of a carrier frequency that is a multipleof said prescribed scan line rate and that is at least as high infrequency as said upper video frequency; means for reversing the phaseof the carrier frequency at said prescribed scan line rate, eachreversal being at a respective instant between successive scan lines, tosupply a folding carrier; a lowpass filter responsive to said videosignal for supplying a low band video signal extending up to a frequencyat least as high as one half said carrier frequency; a highpass filterresponsive to said video signal for supplying a high band video signalextending up from a frequency at least as low as one half said carrierfrequency; means for adaptively deemphasizing said high band videosignal to generate a deemphasized high band video signal; means forgenerating amplitude modulation of said folding carrier in accordancewith said deemphasized high band video signal; means for combining saidlow band video signal with at least the portion of said amplitudemodulation of said folding carrier extending up to a frequency at leastas high as one half said carrier frequency, thereby providing afolded-spectrum video signal.
 41. Apparatus as set forth in claim 40,wherein said means for adaptively deemphasizing said high band videosignal to generate said deemphasized high band video signal comprises: arectifier for generating a rectified response to said high band videosignal; a threshold circuit for responding only to portions of saidrectified response exceeding a predetermined threshold level; a lowpassfilter for generating said deemphasis gain control signal by lowpassfiltering said high band video signal, said deemphasis gain controlsignal being varied in response to the energy level of said high bandvideo signal exceeding said predetermined threshold level; meansresponsive to said deemphasis gain control signal, for generating amultiplier signal by multiplying said high band video signal by saiddeemphasis gain control signal; and means for additively combining saidmultiplied signal with said low band video signal to provide saidmodified full-band video signal.
 42. Apparatus for reversing thespectrum of a video signal descriptive of the scanning of successiveimage frames supplied at a prescribed frame rate, which scanning is doneline-by-line at a prescribed scan line rate, said apparatus comprising:a source of said video signal having a lower video frequency portion andan upper video frequency portion; a generator of a carrier signal of acarrier frequency that is a multiple of said prescribed scan line rateand above said upper video frequency portion; a multiplier having amultiplicand input terminal coupled to receive one of said video signaland said carrier signal and a multiplier input terminal coupled toreceive another of said video signal and said carrier signal, forgenerating a product signal having a first amplitude-modulation sidebandthat is lower in frequency than said carrier frequency and having asecond amplitude-modulation sideband that is higher in frequency thansaid carrier frequency; de-emphasis means for de-emphasizing said uppervideo frequency portion of said video signal and providing said videosignal with deemphasized high frequencies to said multiplier forenabling generation of said product signal; and means for reversing atsaid prescribed scan line rate the phase of the carrier signal receivedby said multiplier, each reversal being at a respective instant betweensuccessive scan lines, thereby enabling conversion of said productsignal into a reversed-spectrum video signal with line-to-line inversionof phase.
 43. Apparatus as set forth in claim 42, wherein saidde-emphasis means comprises: a band-separation filter responding to saidvideo signal for separating said lower video frequency portion and saidupper video frequency portion from said video signal, said lower videofrequency portion comprising only spectral components in alower-frequency band extending up to said carrier frequency, and saidupper video frequency portion comprising only spectral components in anupper-frequency band extending above said carrier frequency; a rectifierfor generating a rectified response to said upper video frequencyportion separated from said video signal; a threshold circuit forresponding only to portions of said rectified response exceeding apredetermined threshold level; a lowpass filter for producing inresponse to the response of said threshold circuit a deemphasis gaincontrol signal dependent on the amount the level of said upper videofrequency portion exceeds said predetermined threshold level; meansresponsive to said deemphasis gain control signal, for generating amultiplier signal by multiplying said upper video frequency portion ofsaid video signal by said gain control signal; and means for additivelycombining said multiplied signal with said lower video frequency portionof said video signal to provide said video signal having saidde-emphasized high frequencies to said multiplier for enablinggeneration of said product signal having said first amplitude-modulationsideband that is lower in frequency than said carrier frequency andhaving said second amplitude-modulation sideband that is higher infrequency than said carrier frequency.
 44. Apparatus as set forth inclaim 43, wherein said band-separation filter comprises: a highpassfilter for highpass filtering said video signal to separate said uppervideo frequency portion thereof; and a lowpass filter for lowpassfiltering said video signal to separate said lower video frequencyportion thereof.
 45. Apparatus as set forth in claim 43, wherein saidmeans responsive to said deemphasis gain control signal for generatingsaid multiplier signal essentially consists of means for subtractingsaid deemphasis gain control signal from a prescribed level. 46.Apparatus as set forth in claim 44, wherein said band-separation filteris a tapped-delay-line filter comprising: a tapped delay line coupled toreceive said video signal, having an odd number of taps for supplyingconsecutively numbered tap signals with increasing increments of delay;a first weighted summation circuit for generating a first weighted sumof odd-numbered ones of said consecutively numbered tap signals, asweighted by weights of a first polarity; a second weighted summationcircuit for generating a second weighted sum of even-numbered ones ofsaid consecutively numbered tap signals, as weighted by weights of firstpolarity; an adder for additively combining said first and secondweighted sums to provide said lower frequency portion of said videosignal; and a subtractor for differentially combining said first andsecond weighted sums to provide said upper frequency portion of saidvideo signal.
 47. Apparatus as set forth in claim 44, wherein saidband-separation filter is a tapped-delay-line filter comprising: atapped delay line coupled to receive said video signal, having an oddnumber of taps for supplying consecutively numbered tap signals withincreasing increments of delay; a first weighted summation circuit forgenerating a first weighted sum of odd-numbered ones of saidconsecutively numbered tap signals, as weighted by weights of a firstpolarity; a second weighted summation circuit for generating a secondweighted sum of even-numbered ones of said consecutively numbered tapsignals, as weighted by weights of said first polarity; a third weightedsummation circuit for generating a third weighted sum of even-numberedones of said consecutively numbered tap signals, as weighted by weightsof a second polarity opposite to said first polarity; a first addercombining said first and second weighted sums to provide said lowerfrequency portion of said video signal; and a second adder combiningsaid first and third weighted sums to provide said upper frequencyportion of said video signal.
 48. Apparatus as set forth in claim 43,wherein said band-separation filter comprises: a linear-phasefinite-impulse-response filter comprising a tapped delay line with anodd number of taps spaced at one sample intervals, and a weightedsummation circuit for weight samples with alternate negative andpositive weights to provide a symmetrical kernel indicative of ahighpass filter response of said video signal; and a subtractor forsubtracting from said video signal said upper video frequency portionseparated from said video signal to provide said lower video frequencyportion.
 49. Apparatus as set forth in claim 42, wherein saidde-emphasis means comprises: means for separating said lower videofrequency portion and said upper video frequency portion from said videosignal; means for rectifying said upper video frequency portionseparated from said video signal to generate a rectified upper videofrequency response; means for baseline clipping said rectified uppervideo frequency response to generate a deemphasis gain control signal;means for generating a multiplier signal having an amplitude that variesin response to said deemphasis gain control signal, growing smaller withincrease in said deemphasis gain control signal and growing larger withdecrease in said deemphasis gain control signal; means for additivelycombining said multiplier signal with said lower video frequency portionseparated from said video signal to provide said video signal havingsaid de-emphasized high frequencies to said multiplier for enablinggeneration of said product signal having said first amplitude-modulationsideband that is lower in frequency than said carrier frequency andhaving said second amplitude-modulation sideband that is higher infrequency than said carrier frequency.
 50. Apparatus as set forth inclaim 49, wherein said means for separating said lower video frequencyportion and said upper video frequency portion from said video signalcomprises: a highpass filter for highpass filtering said video signal togenerate said upper video frequency portion; and a lowpass filter forlowpass filtering said video signal to generate said lower videofrequency portion.
 51. Apparatus as set forth in claim 44, wherein saidband-separation filter is a tapped-delay-line filter comprising: atapped delay line coupled to receive said video signal, having an oddnumber of taps for supplying consecutively numbered tap signals withincreasing increments of delay; a first weighted summation circuit forgenerating a first weighted sum of odd-numbered ones of saidconsecutively numbered tap signals, as weighted by weights of a firstpolarity; a second weighted summation circuit for generating a secondweighted sum of even-numbered ones of said consecutively numbered tapsignals, as weighted by weights of first polarity; an adder foradditively combining said first and second weighted sums to provide saidlower frequency portion of said video signal; and a subtractor fordifferentially combining said first and second weighted sums to providesaid upper frequency portion of said video signal.
 52. Apparatus as setforth in claim 44, wherein said band-separation filter is atapped-delay-line filter comprising: a tapped delay line coupled toreceive said video signal, having an odd number of taps for supplyingconsecutively numbered tap signals with increasing increments of delay;a first weighted summation circuit for generating a first weighted sumof odd-numbered ones of said consecutively numbered tap signals, asweighted by weights of a first polarity; a second weighted summationcircuit for generating a second weighted sum of even-numbered ones ofsaid consecutively numbered tap signals, as weighted by weights of saidfirst polarity; a third weighted summation circuit for generating athird weighted sum of even-numbered ones of said consecutively numberedtap signals, as weighted by weights of a second polarity opposite tosaid first polarity; a first adder combining said first and secondweighted sums to provide said lower frequency portion of said videosignal; and a second adder combining said first and third weighted sumsto provide said upper frequency portion of said video signal.